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SWITCHING REGULATORS
SECTION 3 SWITCHING REGULATORS Walt Kester, Brian Erisman
INTRODUCTION
Virtually all of today's electronic systems require some form of power conversion. The trend toward lower power, portable equipment has driven the technology and the requirement for converting power efficiently. Switchmode power converters, often referred to simply as "switchers", offer a versatile way of achieving this goal. Modern IC switching regulators are small, flexible, and allow either step-up (boost) or step-down (buck) operation. When switcher functions are integrated and include a switch which is part of the basic power converter topology, these ICs are called "switching regulators". When no switches are included in the IC, but the signal for driving an external switch is provided, it is called a "switching regulator controller". Sometimes - usually for higher power levels - the control is not entirely integrated, but other functions to enhance the flexibility of the IC are included instead. In this case the device might be called a "controller" of sorts - perhaps a "feedback controller" if it just generates the feedback signal to the switch modulator. It is important to know what you are getting in your controller, and to know if your switching regulator is really a regulator or is it just the controller function. Also, like switchmode power conversion, linear power conversion and charge pump technology offer both regulators and controllers. So within the field of power conversion, the terms "regulator" and "controller" can have wide meaning. The most basic switcher topologies require only one transistor which is essentially used as a switch, one diode, one inductor, a capacitor across the output, and for practical but not fundamental reasons, another one across the input. A practical converter, however, requires several additional elements, such as a voltage reference, error amplifier, comparator, oscillator, and switch driver, and may also include optional features like current limiting and shutdown capability. Depending on the power level, modern IC switching regulators may integrate the entire converter except for the main magnetic element(s) (usually a single inductor) and the input/output capacitors. Often, a diode, the one which is an essential element of basic switcher topologies, cannot be integrated either. In any case, the complete power conversion for a switcher cannot be as integrated as a linear regulator, for example. The requirement of a magnetic element means that system designers are not inclined to think of switching regulators as simply "drop in" solutions. This presents the challenge to switching regulator manufacturers to provide careful design guidelines, commonly-used application circuits, and plenty of design assistance and product support. As the power levels increase, ICs tend to grow in complexity because it becomes more critical to optimize the control flexibility and precision. Also, since the switches begin to dominate the size of the die, it becomes more cost effective to remove them and integrate only the controller.
3.1
SWITCHING REGULATORS The primary limitations of switching regulators as compared to linear regulators are their output noise, EMI/RFI emissions, and the proper selection of external support components. Although switching regulators do not necessarily require transformers, they do use inductors, and magnetic theory is not generally well understood. However, manufacturers of switching regulators generally offer applications support in this area by offering complete data sheets with recommended parts lists for the external inductor as well as capacitors and switching elements. One unique advantage of switching regulators lies in their ability to convert a given supply voltage with a known voltage range to virtually any given desired output voltage, with no "first order" limitations on efficiency. This is true regardless of whether the output voltage is higher or lower than the input voltage - the same or the opposite polarity. Consider the basic components of a switcher, as stated above. The inductor and capacitor are, ideally, reactive elements which dissipate no power. The transistor is effectively, ideally, a switch in that it is either "on", thus having no voltage dropped across it while current flows through it, or "off", thus having no current flowing through it while there is voltage across it. Since either voltage or current are always zero, the power dissipation is zero, thus, ideally, the switch dissipates no power. Finally, there is the diode, which has a finite voltage drop while current flows through it, and thus dissipates some power. But even that can be substituted with a synchronized switch, called a "synchronous rectifier", so that it ideally dissipates no power either. Switchers also offer the advantage that, since they inherently require a magnetic element, it is often a simple matter to "tap" an extra winding onto that element and, often with just a diode and capacitor, generate a reasonably well regulated additional output. If more outputs are needed, more such taps can be used. Since the tap winding requires no electrical connection, it can be isolated from other circuitry, or made to "float" atop other voltages. Of course, nothing is ideal, and everything has a price. Inductors have resistance, and their magnetic cores are not ideal either, so they dissipate power. Capacitors have resistance, and as current flows in and out of them, they dissipate power, too. Transistors, bipolar or field-effect, are not ideal switches, and have a voltage drop when they are turned on, plus they cannot be switched instantly, and thus dissipate power while they are turning on or off. As we shall soon see, switchers create ripple currents in their input and output capacitors. Those ripple currents create voltage ripple and noise on the converter's input and output due to the resistance, inductance, and finite capacitance of the capacitors used. That is the conducted part of the noise. Then there are often ringing voltages in the converter, parasitic inductances in components and PCB traces, and an inductor which creates a magnetic field which it cannot perfectly contain within its core - all contributors to radiated noise. Noise is an inherent by-product of a switcher and must be controlled by proper component selection, PCB layout, and, if that is not sufficient, additional input or output filtering or shielding.
3.2
SWITCHING REGULATORS
INTEGRATED CIRCUIT SWITCHING REGULATORS
n Advantages: u High Efficiency u Small u Flexible - Step-Up (Boost), Step-Down (Buck), etc. n Disadvantages u Noisy (EMI, RFI, Peak-to-Peak Ripple) u Require External Components (L's, C's) u Designs Can Be Tricky u Higher Total Cost Than Linear Regulators n "Regulators" vs. "Controllers" Figure 3.1
Though switchers can be designed to accommodate a range of input/output conditions, it is generally more costly in non-isolated systems to accommodate a requirement for both voltage step-up and step-down. So generally it is preferable to limit the input/output ranges such that one or the other case can exist, but not both, and then a simpler converter design can be chosen. The concerns of minimizing power dissipation and noise as well as the design complexity and power converter versatility set forth the limitations and challenges for designing switchers, whether with regulators or controllers. The ideal switching regulator shown in Figure 3.2 performs a voltage conversion and input/output energy transfer without loss of power by the use of purely reactive components. Although an actual switching regulator does have internal losses, efficiencies can be quite high, generally greater than 80 to 90%. Conservation of energy applies, so the input power equals the output power. This says that in stepdown (buck) designs, the input current is lower than the output current. On the other hand, in step-up (boost) designs, the input current is greater than the output current. Input currents can therefore be quite high in boost applications, and this should be kept in mind, especially when generating high output voltages from batteries.
3.3
SWITCHING REGULATORS
THE IDEAL SWITCHING REGULATOR
vin + iin LOSSLESS SWITCHING REGULATOR iout vout
Pin
Pout
LOAD
n Pin = Pout n Efficiency = Pout / Pin = 100% n vin · iin n = vout · iout v out i = in vin iout
n Energy Must be Conserved! Figure 3.2
Design engineers unfamiliar with IC switching regulators are sometimes confused by what exactly these devices can do for them. Figure 3.3 summarizes what to expect from a typical IC switching regulator. It should be emphasized that these are typical specifications, and can vary widely, but serve to illustrate some general characteristics. Input voltages may range from 0.8 to beyond 30V, depending on the breakdown voltage of the IC process. Most regulators are available in several output voltage options, 12V, 5V, 3.3V, and 3V are the most common, and some regulators allow the output voltage to be set using external resistors. Output current varies widely, but regulators with internal switches have inherent current handling limitations that controllers (with external switches) do not. Output line and load regulation is typically about 50mV. The output ripple voltage is highly dependent upon the external output capacitor, but with care, can be limited to between 20mV and 100mV peak-to-peak. This ripple is at the switching frequency, which can range from 20kHz to 1MHz. There are also high frequency components in the output current of a switching regulator, but these can be minimized with proper external filtering, layout, and grounding. Efficiency can also vary widely, with up to 95% sometimes being achievable.
3.4
SWITCHING REGULATORS
WHAT TO EXPECT FROM A SWITCHING REGULATOR IC
n Input Voltage Range: 0.8V to 30V n Output Voltage: u "Standard": 12V, 5V, 3.3V, 3V u "Specialized": VID Programmable for Microprocessors u (Some are Adjustable) n Output Current u Up to 1.5A, Using Internal Switches of a Regulator u No Inherent Limitations Using External Switches with a Controller n Output Line / Load Regulation: 50mV, typical n Output Voltage Ripple (peak-peak) : 20mV - 100mV @ Switching Frequency n Switching Frequency: 20kHz - 1MHz n Efficiency: Up to 95%
Figure 3.3
POPULAR APPLICATIONS OF SWITCHING REGULATORS
For equipment which is powered by an AC source, the conversion from AC to DC is generally accomplished with a switcher, except for low-power applications where size and efficiency concerns are outweighed by cost. Then the power conversion may be done with just an AC transformer, some diodes, a capacitor, and a linear regulator. The size issue quickly brings switchers back into the picture as the preferable conversion method as power levels rise up to 10 watts and beyond. Off-line power conversion is heavily dominated by switchers in most modern electronic equipment. Many modern high-power off-line power supply systems use the distributed approach by employing a switcher to generate an intermediate DC voltage which is then distributed to any number of DC/DC converters which can be located near to their respective loads (see Figure 3.4). Although there is the obvious redundancy of converting the power twice, distribution offers some advantages. Since such systems require isolation from the line voltage, only the first converter requires the isolation; all cascaded converters need not be isolated, or at least not to the degree of isolation that the first converter requires. The intermediate DC voltage is usually regulated to less than 60 volts in order to minimize the isolation requirement for the cascaded converters. Its regulation is not critical since it is not a direct output. Since it is typically higher than any of the switching regulator output voltages, the distribution current is substantially less than the sum of the output currents, thereby reducing I2R losses in the system power distribution wiring. This also allows the use of a smaller energy storage capacitor on the intermediate DC supply output. (Recall that the energy stored in a capacitor is ½CV2).
3.5
SWITCHING REGULATORS Power management can be realized by selectively turning on or off the individual DC/DC converters as needed.
POWER DISTRIBUTION USING LINEAR AND SWITCHING REGULATORS
TRADITIONAL USING LINEAR REGULATORS
RECTIFIER AND FILTER AC VN SW REG LINEAR REG V1 SW REG OFF LINE SW REG
DISTRIBUTED USING SWITCHING REGULATORS
V1
AC
VDC < 60V VN
RECTIFIER AND FILTER
LINEAR REG
Figure 3.4
ADVANTAGES OF DISTRIBUTED POWER SYSTEMS USING SWITCHING REGULATORS
n Higher Efficiency with Switching Regulators than Linear Regulators n Use of High Intermediate DC Voltage Minimizes Power Loss due to Wiring Resistance n Flexible (Multiple Output Voltages Easily Obtained) n AC Power Transformer Design Easier (Only One Winding Required, Regulation Not Critical) n Selective Shutdown Techniques Can Be Used for Higher Efficiency n Eliminates Safety Isolation Requirements for DC/DC Converters Figure 3.5 Batteries are the primary power source in much of today's consumer and communications equipment. Such systems may require one or several voltages, and they may be less or greater than the battery voltage. Since a battery is a selfcontained power source, power converters seldom require isolation. Often, then, the basic switcher topologies are used, and a wide variety of switching regulators are 3.6
SWITCHING REGULATORS available to fill many of the applications. Maximum power levels for these regulators typically can range up from as low as tens of milliwatts to several watts. Efficiency is often of great importance, as it is a factor in determining battery life which, in turn, affects practicality and cost of ownership. Often of even greater importance, though often confused with efficiency, is quiescent power dissipation when operating at a small fraction of the maximum rated load (e.g., standby mode). For electronic equipment which must remain under power in order to retain data storage or minimal monitoring functions, but is otherwise shut down most of the time, the quiescent dissipation is the largest determinant of battery life. Although efficiency may indicate power consumption for a specific light load condition, it is not the most useful way to address the concern. For example, if there is no load on the converter output, the efficiency will be zero no matter how optimal the converter, and one could not distinguish a well power-managed converter from a poorly managed one by such a specification. The concern of managing power effectively from no load to full load has driven much of the technology which has been and still is emerging from today's switching regulators and controllers. Effective power management, as well as reliable power conversion, is often a substantial factor of quality or noteworthy distinction in a wide variety of equipment. The limitations and cost of batteries are such that consumers place a value on not having to replace them more often than necessary, and that is certainly a goal for effective power conversion solutions.
TYPICAL APPLICATION OF A BOOST REGULATOR IN BATTERY OPERATED EQUIPMENT
STEP-UP (BOOST) SWITCHING REGULATOR + VBATTERY LOAD VOUT > VBATTERY
Figure 3.6
3.7
SWITCHING REGULATORS
INDUCTOR AND CAPACITOR FUNDAMENTALS
In order to understand switching regulators, the fundamental energy storage capabilities of inductors and capacitors must be fully understood. When a voltage is applied to an ideal inductor (see Figure 3.7), the current builds up linearly over time at a rate equal to V/L, where V is the applied voltage, and L is the value of the inductance. This energy is stored in the inductor's magnetic field, and if the switch is opened, the magnetic field collapses, and the inductor voltage goes to a large instantaneous value until the field has fully collapsed.
INDUCTOR AND CAPACITOR FUNDAMENTALS
i
+ V
L
I
+ v -
C
V=L
di dt
di V = dt L
v
I= C
dv dt
dv I = dt C
i
t Current Does Not Change Instantaneously
0
0
t
Voltage Does Not Change Instantaneously
Figure 3.7 When a current is applied to an ideal capacitor, the capacitor is gradually charged, and the voltage builds up linearly over time at a rate equal to I/C, where I is the applied current, and C is the value of the capacitance. Note that the voltage across an ideal capacitor cannot change instantaneously. Of course, there is no such thing as an ideal inductor or capacitor. Real inductors have stray winding capacitance, series resistance, and can saturate for large currents. Real capacitors have series resistance and inductance and may break down under large voltages. Nevertheless, the fundamentals of the ideal inductor and capacitor are critical in understanding the operation of switching regulators. An inductor can be used to transfer energy between two voltage sources as shown in Figure 3.8. While energy transfer could occur between two voltage sources with a resistor connected between them, the energy transfer would be inefficient due to the power loss in the resistor, and the energy could only be transferred from the higher to the lower value source. In contrast, an inductor ideally returns all the energy that
3.8
SWITCHING REGULATORS is stored in it, and with the use of properly configured switches, the energy can flow from any one source to another, regardless of their respective values and polarities.
ENERGY TRANSFER USING AN INDUCTOR
i1 + V1 L iL + V2 i2
iL 0 i1 0
V1 L
IPEAK t1 IPEAK
E=
V - 2 L
(SLOPE) t
t2
1 L · I PEAK 2 2
t IPEAK
i2 0 t
Figure 3.8 When the switches are initially placed in the position shown, the voltage V1 is applied to the inductor, and the inductor current builds up at a rate equal to V1/L. The peak value of the inductor current at the end of the interval t1 is V IPEAK = 1 · t1 . L The average power transferred to the inductor during the interval t1 is PAVG = 1 I PEAK · V1 . 2
The energy transferred during the interval t1 is E = PAVG · t1 = 1 I PEAK · V1 · t1 . 2
Solving the first equation for t1 and substituting into the last equation yields E= 1 L · I PEAK 2 . 2
3.9
SWITCHING REGULATORS When the switch positions are reversed, the inductor current continues to flow into the load voltage V2, and the inductor current decreases at a rate V2/L. At the end of the interval t2, the inductor current has decreased to zero, and the energy has been transferred into the load. The figure shows the current waveforms for the inductor, the input current i1, and the output current i2. The ideal inductor dissipates no power, so there is no power loss in this transfer, assuming ideal circuit elements. This fundamental method of energy transfer forms the basis for all switching regulators.
IDEAL STEP-DOWN (BUCK) CONVERTER
The fundamental circuit for an ideal step-down (buck) converter is shown in Figure 3.9. The actual integrated circuit switching regulator contains the switch control circuit and may or may not include the switch (depending upon the output current requirement). The inductor, diode, and load bypass capacitor are external.
BASIC STEP-DOWN (BUCK) CONVERTER
ERROR AMPLIFIER AND SWITCH CONTROL CIRCUIT
SENSE
SW L
+
C D
LOAD
f= SW ON ton
1 t on + t off
SW OFF
toff
Figure 3.9 The output voltage is sensed and then regulated by the switch control circuit. There are several methods for controlling the switch, but for now assume that the switch is controlled by a pulse width modulator (PWM) operating at a fixed frequency, f. The actual waveforms associated with the buck converter are shown in Figure 3.10. When the switch is on, the voltage VINVOUT appears across the inductor, and the inductor current increases with a slope equal to (VINVOUT)/L (see Figure 3.10B). When the switch turns off, current continues to flow through the inductor and into the load (remember that the current cannot change instantaneously in an inductor), with the ideal diode providing the return current path. The voltage across the inductor is now VOUT, but the polarity has reversed. Therefore, the inductor 3.10
SWITCHING REGULATORS current decreases with a slope equal to VOUT/L. Note that the inductor current is equal to the output current in a buck converter. The diode and switch currents are shown in Figures 3.10C and 3.10D, respectively, and the inductor current is the sum of these waveforms. Also note by inspection that the instantaneous input current equals the switch current. Note, however, that the average input current is less than the average output current. In a practical regulator, both the switch and the diode have voltage drops across them during their conduction which creates internal power dissipation and a loss of efficiency, but these voltages will be neglected for now. It is also assumed that the output capacitor, C, is large enough so that the output voltage does not change significantly during the switch on or off times.
BASIC STEP-DOWN (BUCK) CONVERTER WAVEFORMS
iIN = iSW IIN VIN SW + C iD D LOAD IOUT iD 0 Lower Case = Instantaneous Value Upper Case = Average Value iIN = iSW 0 vD L iL = iOUT IOUT VOUT iL = iOUT 0 vD VIN ton 0 IOUT toff ton
A
B
VIN - VOUT L - VOUT L
(SLOPES)
C
IOUT IIN
D
Figure 3.10 There are several important things to note about these waveforms. The most important is that ideal components have been assumed, i.e., the input voltage source has zero impedance, the switch has zero on-resistance and zero turn-on and turn-off times. It is also assumed that the inductor does not saturate and that the diode is ideal with no forward drop. Also note that the output current is continuous, while the input current is pulsating. Obviously, this has implications regarding input and output filtering. If one is concerned about the voltage ripple created on the power source which supplies a buck converter, the input filter capacitor (not shown) is generally more critical that the output capacitor with respect to ESR/ESL.
3.11
SWITCHING REGULATORS If a steady-state condition exists (see Figure 3.11), the basic relationship between the input and output voltage may be derived by inspecting the inductor current waveform and writing: VIN - VOUT V · t on = OUT · t off . L L Solving for VOUT: VOUT = VIN · t on = VIN · D , t on + t off
where D is the switch duty ratio (more commonly called duty cycle), defined as the ratio of the switch on-time (ton) to the total switch cycle time (ton + toff). This is the classic equation relating input and output voltage in a buck converter which is operating with continuous inductor current, defined by the fact that the inductor current never goes to zero.
INPUT/OUTPUT RELATIONSHIP FOR BUCK CONVERTER
ton toff ton
IOUT iL = iOUT 0
VIN - VOUT L - VOUT L
n Write by Inspection from Inductor/Output Current Waveforms: n
VIN - VOUT V · t on = OUT · t off L L
n Rearrange and Solve for VOUT: n VOUT = VIN · t
t on = VIN · D on + t off
Figure 3.11 Notice that this relationship is independent of the inductor value L as well as the switching frequency 1/(ton + toff) and the load current. Decreasing the inductor value, however, will result in a larger peak-to-peak output ripple current, while increasing the value results in smaller ripple. There are many other tradeoffs involved in selecting the inductor, and these will be discussed in a later section.
3.12
SWITCHING REGULATORS In this simple model, line and load regulation (of the output voltage) is achieved by varying the duty cycle using a pulse width modulator (PWM) operating at a fixed frequency, f. The PWM is in turn controlled by an error amplifier - an amplifier which amplifies the "error" between the measured output voltage and a reference voltage. As the input voltage increases, the duty cycle decreases; and as the input voltage decreases, the duty cycle increases. Note that while the average inductor current changes proportionally to the output current, the duty cycle does not change. Only dynamic changes in the duty cycle are required to modulate the inductor current to the desired level; then the duty cycle returns to its steady state value. In a practical converter, the duty cycle might increase slightly with load current to counter the increase in voltage drops in the circuit, but would otherwise follow the ideal model. This discussion so far has assumed the regulator is in the continuous-mode of operation, defined by the fact that the inductor current never goes to zero. If, however, the output load current is decreased, there comes a point where the inductor current will go to zero between cycles, and the inductor current is said to be discontinuous. It is necessary to understand this operating mode as well, since many switchers must supply a wide dynamic range of output current, where this phenomenon is unavoidable. Waveforms for discontinuous operation are shown in Figure 3.12.
BUCK CONVERTER WAVEFORMS DISCONTINUOUS MODE
iIN = iSW IIN VIN SW + C iD D LOAD iD 0 iIN = iSW 0 vD L iL = iOUT IOUT VOUT vD VIN VOUT 0 iL = iOUT 0 ton toff
A
ton
IOUT
B
C
Lower Case = Instantaneous Value Upper Case = Average Value
IIN
D
Figure 3.12 Behavior during the switch on-time is identical to that of the continuous mode of operation. However, during the switch off-time, there are two regions of unique behavior. First, the inductor current ramps down at the same rate as it does during continuous mode, but then the inductor current goes to zero. When it reaches zero, the current tries to reverse but cannot find a path through the diode any longer. So the voltage on the input side of the inductor (same as the diode and switch junction)
3.13
SWITCHING REGULATORS jumps up to VOUT such that the inductor has no voltage across it, and the current can remain at zero. Because the impedance at diode node (vD) is high, ringing occurs due to the inductor, L, resonating with the stray capacitance which is the sum of the diode capacitance, CD, and the switch capacitance, CSW. The oscillation is damped by stray resistances in the circuit, and occurs at a frequency given by fo = 1 2 L (C D + CSW ) .
A circuit devoted simply to dampening resonances via power dissipation is called a snubber. If the ringing generates EMI/RFI problems, it may be damped with a suitable RC snubber. However, this will cause additional power dissipation and reduced efficiency. If the load current of a standard buck converter is low enough, the inductor current becomes discontinuous. The current at which this occurs can be calculated by observing the waveform shown in Figure 3.13. This waveform is drawn showing the inductor current going to exactly zero at the end of the switch off-time. Under these conditions, the average output current is IOUT = IPEAK/2. We have already shown that the peak inductor current is I PEAK = VIN - VOUT · t on . L
Thus, discontinuous operation will occur if IOUT < VIN - VOUT · t on . 2L
However, VOUT and VIN are related by: VOUT = VIN · D = VIN · Solving for ton: t on = V VOUT 1 · ( t on + t off ) = OUT · . VIN VIN f t on . t on + t off
3.14
SWITCHING REGULATORS Substituting this value for ton into the previous equation for IOUT: V VOUT 1 - OUT V IN IOUT < . 2Lf
(Criteria for discontinuous operation buck converter)
BUCK CONVERTER POINT OF DISCONTINUOUS OPERATION
INDUCTOR CURRENT AND OUTPUT CURRENT
VIN - VOUT L
IPEAK
- VOUT L
IOUT ton toff
0
DISCONTINUOUS MODE IF:
IOUT < V - VOUT 1 I PEAK = IN · t on 2 2L
V VOUT 1 - OUT VIN IOUT < , 2Lf
f=
1 t on + t off
Figure 3.13
IDEAL STEP-UP (BOOST) CONVERTER
The basic step-up (boost) converter circuit is shown in Figure 3.14. During the switch on-time, the current builds up in the inductor. When the switch is opened, the energy stored in the inductor is transferred to the load through the diode. The actual waveforms associated with the boost converter are shown in Figure 3.15. When the switch is on, the voltage VIN appears across the inductor, and the inductor current increases at a rate equal to VIN/L. When the switch is opened, a voltage equal to VOUT VIN appears across the inductor, current is supplied to the load, and the current decays at a rate equal to (VOUT VIN)/L. The inductor current waveform is shown in Figure 3.15B.
3.15
SWITCHING REGULATORS
BASIC STEP-UP (BOOST) CONVERTER
ERROR AMPLIFIER AND SWITCH CONTROL CIRCUIT
SENSE
L
+
D SW C
LOAD
f= SW ON ton
1 t on + t off
SW OFF
toff
Figure 3.14
BASIC STEP-UP (BOOST) CONVERTER WAVEFORMS
iIN = iL IIN VIN L + iSW SW LOAD IIN iSW 0 Lower Case = Instantaneous Value Upper Case = Average Value iD = iOUT 0 vD D C iD = iOUT IOUT VOUT iIN= iL 0 VOUT vSW 0 IIN VIN L VIN - VOUT L ton toff ton
A B
(SLOPES)
C
IIN IOUT
D
Figure 3.15
3.16
SWITCHING REGULATORS
Note that in the boost converter, the input current is continuous, while the output current (Figure 3.15D) is pulsating. This implies that filtering the output of a boost converter is more difficult than that of a buck converter. (Refer back to the previous discussion of buck converters). Also note that the input current is the sum of the switch and diode current. If a steady-state condition exists (see Figure 3.16), the basic relationship between the input and output voltage may be derived by inspecting the inductor current waveform and writing: VIN V - VIN · t on = OUT · t off . L L Solving for VOUT: 1 t + t off = VIN · . VOUT = VIN · on t off 1-D
INPUT/OUTPUT RELATIONSHIP FOR BOOST CONVERTER
ton toff ton
iL = iIN 0
VIN L
VIN - VOUT L
IOUT
n Write by Inspection from Inductor/Input Current Waveforms: n
VIN V - VIN · t on = OUT · t off L L
n Rearrange and Solve for VOUT: n
t + t off 1 VOUT = VIN · on = VIN · t off 1- D
Figure 3.16
3.17
SWITCHING REGULATORS
This discussion so far has assumed the boost converter is in the continuous-mode of operation, defined by the fact that the inductor current never goes to zero. If, however, the output load current is decreased, there comes a point where the inductor current will go to zero between cycles, and the inductor current is said to be discontinuous. It is necessary to understand this operating mode as well, since many switchers must supply a wide dynamic range of output current, where this phenomenon is unavoidable. Discontinuous operation for the boost converter is similar to that of the buck converter. Figure 3.17 shows the waveforms. Note that when the inductor current goes to zero, ringing occurs at the switch node at a frequency fo given by: 1 2 L ( CD + CSW )
fo =
.
BOOST CONVERTER WAVEFORMS DISCONTINUOUS MODE
iIN = iL IIN VIN vSW D C iSW SW LOAD iSW 0 iIN = iL 0 iD = iOUT IOUT VOUT vSW VOUT VIN 0 ton toff IIN ton
A
+
L
B
C
Lower Case = Instantaneous Value Upper Case = Average Value
iD = iOUT 0
IOUT
D
Figure 3.17 The inductor, L, resonates with the stray switch capacitance and diode capacitance, CSW + CD as in the case of the buck converter. The ringing is dampened by circuit resistances, and, if needed, a snubber. The current at which a boost converter becomes discontinuous can be derived by observing the inductor current (same as input current) waveform of Figure 3.18.
3.18
SWITCHING REGULATORS
BOOST CONVERTER POINT OF DISCONTINUOUS OPERATION
INDUCTOR CURRENT AND INPUT CURRENT
VIN L IIN ton toff IPEAK
VIN - VOUT L
0
DISCONTINUOUS MODE IF:
IIN < 1 V - VIN · t off I PEAK = OUT 2 2L
f= 1 t on + t off
V 2 ( VOUT - VIN) IOUT < IN , VOUT 2 · 2Lf
Figure 3.18 The average input current at the point of discontinuous operation is IIN = IPEAK/2. Discontinuous operation will occur if IIN < IPEAK/2. However, I V - VIN I IN = PEAK = OUT · t off . 2 2L Also,
VIN · I IN = VOUT · I OUT , and therefore
IOUT = However,
( VOUT - VIN ) VIN VIN · I IN = · · t off . VOUT VOUT 2L
VOUT 1 = = VIN 1-D
1 t + t off = on . t on t off 1- t on + t off 3.19
SWITCHING REGULATORS
Solving for toff: VIN V (t on + t off ) = f · VIN . VOUT OUT
t off =
Substituting this value for toff into the previous expression for IOUT, the criteria for discontinuous operation of a boost converter is established: VIN 2 ( VOUT - VIN ) . VOUT 2 · 2Lf
IOUT <
(Criteria for discontinuous operation boost converter).
The basic buck and boost converter circuits can work equally well for negative inputs and outputs as shown in Figure 3.19. Note that the only difference is that the polarities of the input voltage and the diode have been reversed. In practice, however, not many IC buck and boost regulators or controllers will work with negative inputs. In some cases, external circuitry can be added in order to handle negative inputs and outputs. Rarely are regulators or controllers designed specifically for negative inputs or outputs. In any case, data sheets for the specific ICs will indicate the degree of flexibility allowed.
NEGATIVE IN, NEGATIVE OUT BUCK AND BOOST CONVERTERS
VIN SW L C D + SW LOAD + + VOUT VIN L D C + LOAD VOUT
BUCK Figure 3.19
BOOST
3.20
SWITCHING REGULATORS
BUCK-BOOST TOPOLOGIES
The simple buck converter can only produce an output voltage which is less than the input voltage, while the simple boost converter can only produce an output voltage greater than the input voltage. There are many applications where more flexibility is required. This is especially true in battery powered applications, where the fully charged battery voltage starts out greater than the desired output (the converter must operate in the buck mode), but as the battery discharges, its voltage becomes less than the desired output (the converter must then operate in the boost mode). A buck-boost converter is capable of producing an output voltage which is either greater than or less than the absolute value of the input voltage. A simple buckboost converter topology is shown in Figure 3.20. The input voltage is positive, and the output voltage is negative. When the switch is on, the inductor current builds up. When the switch is opened, the inductor supplies current to the load through the diode. Obviously, this circuit can be modified for a negative input and a positive output by reversing the polarity of the diode.
BUCK-BOOST CONVERTER #1, +VIN, -VOUT
VIN SW VOUT (NEGATIVE)
+
D L C +
LOAD
The Absolute Value of the Output Can Be Less Than Or Greater Than the Absolute Value of the Input
Figure 3.20
3.21
SWITCHING REGULATORS
A second buck-boost converter topology is shown in Figure 3.21. This circuit allows both the input and output voltage to be positive. When the switches are closed, the inductor current builds up. When the switches open, the inductor current is supplied to the load through the current path provided by D1 and D2. A fundamental disadvantage to this circuit is that it requires two switches and two diodes. As in the previous circuits, the polarities of the diodes may be reversed to handle negative input and output voltages.
BUCK-BOOST CONVERTER #2 +VIN, +VOUT
VIN SW1 L D1 SW2 D2 C VOUT (POSITIVE)
+
+
LOAD
The Absolute Value of the Output Can Be Less Than Or Greater Than the Absolute Value of the Input
Figure 3.21
Another way to accomplish the buck-boost function is to cascade two switching regulators; a boost regulator followed by a buck regulator as shown in Figure 3.22. The example shows some practical voltages in a battery-operated system. The input from the four AA cells can range from 6V (charged) to about 3.5V (discharged). The intermediate voltage output of the boost converter is 8V, which is always greater than the input voltage. The buck regulator generates the desired 5V from the 8V intermediate voltage. The total efficiency of the combination is the product of the individual efficiencies of each regulator, and can be greater than 85% with careful design. An alternate topology is use a buck regulator followed by a boost regulator. This approach, however, has the disadvantage of pulsating currents on both the input and output and a higher current at the intermediate voltage output.
3.22
SWITCHING REGULATORS
CASCADED BUCK-BOOST REGULATORS (EXAMPLE VOLTAGES)
VIN, 4 AA CELLS 3.5 - 6V INTERMEDIATE VOLTAGE VOUT 8V 5V
BOOST REGULATOR
BUCK REGULATOR
+
Figure 3.22
OTHER NON-ISOLATED SWITCHER TOPOLOGIES
The coupled-inductor single-ended primary inductance converter (SEPIC) topology is shown in Figure 3.23. This converter uses a transformer with the addition of capacitor CC which couples additional energy to the load. If the turns ratio (N = the ratio of the number of primary turns to the number of secondary turns) of the transformer in the SEPIC converter is 1:1, the capacitor serves only to recover the energy in the leakage inductance (i.e., that energy which is not perfectly coupled between the windings) and delivering it to the load. In that case, the relationship between input and output voltage is given by D . 1- D
VOUT = VIN ·
For non-unity turns ratios the input/output relationship is highly nonlinear due to transfer of energy occurring via both the coupling between the windings and the capacitor CC. For that reason, it is not analyzed here.
3.23
SWITCHING REGULATORS
SINGLE-ENDED PRIMARY INDUCTANCE CONVERTER (SEPIC)
VIN N:1 VOUT
+
CC C
LOAD
Figure 3.23 This converter topology often makes an excellent choice in non-isolated batterypowered systems for providing both the ability to step up or down the voltage, and, unlike the boost converter, the ability to have zero voltage at the output when desired. The Zeta and Cük converters, not shown, are two examples of non-isolated converters which require capacitors to deliver energy from input to output, i.e., rather than just to store energy or deliver only recovered leakage energy, as the SEPIC can be configured via a 1:1 turns ratio. Because capacitors capable of delivering energy efficiently in such converters tend to be bulky and expensive, these converters are not frequently used.
ISOLATED SWITCHING REGULATOR TOPOLOGIES
The switching regulators discussed so far have direct galvanic connections between the input and output. Transformers can be used to supply galvanic isolation as well as allowing the buck-boost function to be easily performed. However, adding a transformer to the circuit creates a more complicated and expensive design as well as increasing the physical size. The basic flyback buck-boost converter circuit is shown in Figure 3.24. It is derived from the buck-boost converter topology. When the switch is on, the current builds up in the primary of the transformer. When the switch is opened, the current reverts to the secondary winding and flows through the diode and into the load. The relationship between the input and output voltage is determined by the turns ratio, N, and the duty cycle, D, per the following equation: VOUT = VIN D · . N 1- D
3.24
SWITCHING REGULATORS
A disadvantage of the flyback converter is the high energy which must be stored in the transformer in the form of DC current in the windings. This requires larger cores than would be necessary with pure AC in the windings.
ISOLATED TOPOLOGY: FLYBACK CONVERTER
N:1
+ VIN
D C LOAD VOUT
SW
(BUCK-BOOST DERIVED) V D VOUT = IN · N 1- D D = Duty Cycle Figure 3.24 The basic forward converter topology is shown in Figure 3.25. It is derived from the buck converter. This topology avoids the problem of large stored energy in the transformer core. However, the circuit is more complex and requires an additional magnetic element (a transformer), an inductor, an additional transformer winding, plus three diodes. When the switch is on, current builds up in the primary winding and also in the secondary winding, where it is transferred to the load through diode D1. When the switch is on, the current in the inductor flows out of D1 from the transformer and is reflected back to the primary winding according to the turns ratio. Additionally, the current due to the input voltage applied across the primary inductance, called the magnetizing current, flows in the primary winding. When the switch is opened, the current in the inductor continues to flow through the load via the return path provided by diode D2. The load current is no longer reflected into the transformer, but the magnetizing current induced in the primary still requires a return path so that the transformer can be reset. Hence the extra reset winding and diode are needed. The relationship between the input and output voltage is given by: VOUT = VIN · D. N
3.25
SWITCHING REGULATORS
ISOLATED TOPOLOGY: FORWARD CONVERTER
N:1 D1 + VIN D2 C D3 SW LOAD VOUT L
(BUCK DERIVED)
VOUT = VIN ·D N
D = Duty Cycle Figure 3.25 There are many other possible isolated switching regulator topologies which use transformers, however, the balance of this section will focus on non-isolated topologies because of their wider application in portable and distributed power systems.
SWITCH MODULATION TECHNIQUES
Important keys to understanding switching regulators are the various methods used to control the switch. For simplicity of analysis, the examples previously discussed used a simple fixed-frequency pulse width modulation (PWM) technique. There can be two other standard variations of the PWM technique: variable frequency constant on-time, and variable frequency constant off-time. In the case of a buck converter, using a variable frequency constant off-time ensures that the peak-to-peak output ripple current (also the inductor current) remains constant as the input voltage varies. This is illustrated in Figure 3.26, where the output current is shown for two conditions of input voltage. Note that as the input voltage increases, the slope during the on-time increases, but the on-time decreases, thereby causing the frequency to increase. Constant off-time control schemes are popular for buck converters where a wide input voltage range must be accomodated. The ADP1147 family implements this switch modulation technique.
3.26
SWITCHING REGULATORS
CONTROL OF BUCK CONVERTER USING CONSTANT OFF-TIME, VARIABLE FREQUENCY PWM
VIN SW + C D LOAD iL = iOUT L VOUT
iL = iOUT WAVEFORMS
VIN - VOUT L
- VOUT L
LARGER VIN
- VOUT L
CONSTANT PEAK-TO-PEAK RIPPLE
Figure 3.26 In the case of a boost converter, however, neither input ramp slopes nor output ramp slopes are solely a function of the output voltage (see Figure 3.15), so there is no inherent advantage in the variable frequency constant off-time modulation method with respect to maintaining constant output ripple current. Still, that modulation method tends to allow for less ripple current variation than does fixed frequency, so it is often used. In the case where very low duty cycles are needed, e.g., under short circuit conditions, sometimes the limitation of a minimum achievable duty cycle is encountered. In such cases, in order to maintain a steady-state condition and prevent runaway of the switch current, a pulse skipping function must be implemented. This might take the form of a current monitoring circuit which detects that the switch current is too high to turn the switch on and ramp the current up any higher. So either a fixed frequency cycle is skipped without turning on the switch, or the off-time is extended in some way to delay the turn-on. The pulse skipping technique for a fixed frequency controller can be applied even to operation at normal duty cycles. Such a switch modulation technique is then referred to as pulse burst modulation (PBM). At its simplest, this technique simply gates a fixed frequency, fixed duty cycle oscillator to be applied to the switch or not. The duty cycle of the oscillator sets the maximum achievable duty cycle for the converter, and smaller duty cycles are achieved over an average of a multiplicity of pulses by skipping oscillator cycles. This switch modulation method accompanies a simple control method of using a hysteretic comparator to monitor the output voltage versus a reference and decide whether to use the oscillator to turn on the switch for that cycle or not. The hysteresis of the comparator tends to give rise to several cycles of switching followed by several cycles of not switching. Hence, the 3.27
SWITCHING REGULATORS resulting switching signal is characterized by pulses which tend to come in bursts hence the name for the modulation technique. There are at least two inherent fundamental drawbacks of the PBM switch modulation technique. First, the constant variation of the duty cycle between zero and maximum produces high ripple currents and accompanying losses. Second, there is an inherent generation of subharmonic frequencies with respect to the oscillator frequency. This means that the noise spectrum is not well controlled, and often audible frequencies can be produced. This is often apparent in higher power converters which use pulse skipping to maintain short-circuit current control. An audible noise can often be heard under such a condition, due to the large magnetics acting like speaker coils. For these reasons, PBM is seldom used at power levels above ~10 Watts. But for its simplicity, it is often preferred below that power level, but above a power level or with a power conversion requirement where charge pumps are not well suited.
CONTROL TECHNIQUES
Though often confused with or used in conjunction with discussing the switch modulation technique, the control technique refers to what parameters of operation are used and how they are used to control the modulation of the switch. The specific way in which the switch is modulated can be thought of separately, and was just presented in the previous section. In circuits using PBM for switch modulation, the control technique typically used is a voltage-mode hysteretic control. In this implementation the switch is controlled by monitoring the output voltage and modulating the switch such that the output voltage oscillates between two hysteretic limits. The ADP3000 switching regulator is an example of a regulator which combines these modulation and control techniques. The most basic control technique for use with PWM is voltage-mode (VM) control (see Figure 3.27). Here, the output voltage is the only parameter used to determine how the switch will be modulated. An error amplifier (first mentioned in the Buck Converter section) monitors the output voltage, its error is amplified with the required frequency compensation for maintaining stability of the control loop, and the switch is modulated directly in accordance with that amplifier output. The output voltage is divided down by a ratio-matched resistor divider and drives one input of an amplifier, G. A precision reference voltage (VREF) is applied to the other input of the amplifier. The output of the amplifier in turn controls the duty cycle of the PWM. It is important to note that the resistor divider, amplifier, and reference are actually part of the switching regulator IC, but are shown externally in the diagram for clarity. The output voltage is set by the resistor divider ratio and the reference voltage: R2 VOUT = VREF 1 + . R1
3.28
SWITCHING REGULATORS The internal resistor ratios and the reference voltage are set to produce standard output voltage options such as 12V, 5V, 3.3V, or 3V. In some regulators, the resistor divider can be external, allowing the output voltage to be adjusted.
VOLTAGE FEEDBACK FOR PWM CONTROL
VIN TO PWM VOUT
+
SWITCHING REG. IC, INDUCTOR, DIODE
LOAD
R2
R1
G
VREF NOTE: RESISTORS, AMPLIFIER, AND VREF INCLUDED IN SWITCHING REGULATOR IC
Figure 3.27 A simple modification of VM control is voltage feedforward. This technique adjusts the duty cycle automatically as the input voltage changes so that the feedback loop does not have to make an adjustment (or as much of an adjustment). Voltage feedforward can even be used in the simple PBM regulators. Feedforward is especially useful in applications where the input voltage can change suddenly or, perhaps due to current limit protection limitations, it is desirable to limit the maximum duty cycle to lower levels when the input voltage is higher. In switchers, the VM control loop needs to be compensated to provide stability, considering that the voltage being controlled by the modulator is the average voltage produced at the switched node, whereas the actual output voltage is filtered through the switcher's LC filter. The phase shift produced by the filter can make it difficult to produce a control loop with a fast response time. A popular way to circumvent the problem produced by the LC filter phase shift is to use current-mode (CM) control as shown in Figure 3.28. In current-mode control, it is still desirable, of course, to regulate the output voltage. Thus, an error amplifier (G1) is still required. However, the switch modulation is no longer controlled directly by the error amplifier. Instead, the inductor current is sensed, amplified by G2, and used to modulate the switch in accordance with the command signal from the [output voltage] error amplifier. It should be noted that the divider network, VREF, G1 and G2 are usually part of the IC switching regulator itself, rather than external as shown in the simplified diagram.
3.29
SWITCHING REGULATORS
CURRENT FEEDBACK FOR PWM CONTROL
TO PWM VIN
G2 iOUT RSENSE R2 VOUT
+
SWITCHING REG. IC, INDUCTOR, DIODE
LOAD
R1
G1
VREF NOTE: RESISTORS, AMPLIFIERS, AND VREF INCLUDED IN SWITCHING REGULATOR IC
Figure 3.28
The CM control system uses feedback from both the output voltage and output current. Recall that at the beginning of each PWM cycle, the switch turns on, and the inductor current begins to rise. The inductor current develops a voltage across the small sense resistor, RSENSE, which is amplified by G2 and fed back to the PWM controller to turn off the switch. The output voltage, sensed by amplifier G1 and also fed back to the PWM controller, sets the level at which the peak inductor current will terminate the switch on-time. Since it is inductor current that turns off the switch (and thereby sets the duty cycle) this method is commonly referred to as current-mode control, even though there are actually two feedback control loops: the fast responding current loop, and the slower responding output voltage loop. Note that inductor current is being controlled on a pulse-by-pulse basis, which simplifies protection against switch over-current and inductor saturation conditions. In essence, then, in CM control, rather than controlling the average voltage which is applied to the LC filter as in VM control, the inductor current is controlled directly on a cycle-by-cycle basis. The only phase shift remaining between the inductor current and the output voltage is that produced by the impedance of the output capacitor(s). The correspondingly lower phase shift in the output filter allows the loop response to be faster while still remaining stable. Also, instantaneous changes in input voltage are immediately reflected in the inductor current, which provides excellent line transient response. The obvious disadvantage of CM control is the requirement of sensing current and, if needed, an additional amplifier. With increasingly higher performance requirements in modern electronic equipment, the performance advantage of CM control typically outweighs the cost of 3.30
SWITCHING REGULATORS implementation. Also, some sort of current limit protection is often required, whatever the control technique. Thus it tends to be necessary to implement some sort of current sensing even in VM-controlled systems. Now even though we speak of a CM controller as essentially controlling the inductor current, more often than not the switch current is controlled instead, since it is more easily sensed (especially in a switching regulator) and it is a representation of the inductor current for at least the on-time portion of the switching cycle. Rather than actually controlling the average switch current, which is not the same as the average inductor current anyway, it is often simpler to control the peak current which is the same for both the switch and the inductor in all the basic topologies. The error between the average inductor current and the peak inductor current produces a non-linearity within the control loop. In most systems, that is not a problem. In other systems, a more precise current control is needed, and in such a case, the inductor current is sensed directly and amplified and frequencycompensated for the best response. Other control variations are possible, including valley rather than peak control, hysteretic current control, and even charge control - a technique whereby the integral of the inductor current (i.e., charge) is controlled. That eliminates even the phase shift of the output capacitance from the loop, but presents the problem that instantaneous current is not controlled, and therefore short-circuit protection is not inherent in the system. All techniques offer various advantages and disadvantages. Usually the best tradeoff between performance and cost/simplicity is peak-current control - as used by the ADP1147 family. This family also uses the current-sense output to control a sleep, or power saving mode of operation to maintain high efficiency for low output currents.
GATED OSCILLATOR (PULSE BURST MODULATION) CONTROL EXAMPLE
All of the PWM techniques discussed thus far require some degree of feedback loop compensation. This can be especially tricky for boost converters, where there is more phase shift between the switch and the output voltage. As previously mentioned, a technique which requires no feedback compensation uses a fixed frequency gated oscillator as the switch control (see Figure 3.29). This method is often (incorrectly) referred to as the Pulse Frequency Modulation (PFM) mode, but is more correctly called pulse burst modulation (PBM) or gated-oscillator control. The output voltage (VOUT) is divided by the resistive divider (R1 and R2) and compared against a reference voltage, VREF. The comparator hysteresis is required for stability and also affects the output voltage ripple. When the resistor divider output voltage drops below the comparator threshold (VREF minus the hysteresis voltage), the comparator starts the gated oscillator. The switcher begins switching again which then causes the output voltage to increase until the comparator threshold is reached (VREF plus the hysteresis voltage), at which time the oscillator is turned off. When the oscillator is off, quiescent current drops to a very low value
3.31
SWITCHING REGULATORS (for example, 95µA in the ADP1073) making PBM controllers very suitable for battery-powered applications.
SWITCH CONTROL USING GATED OSCILLATOR (PULSE BURST MODULATION, PBM)
FIXED FREQUENCY GATED OSCILLATOR
VOUT COMPARATOR WITH HYSTERESIS ON/OFF
SWITCH CONTROL VIN
+
SWITCHING REG. IC, INDUCTOR, DIODE
LOAD
R2
R1
VREF NOTE: RESISTORS, AMPLIFIER, OSCILLATOR AND VREF INCLUDED IN SWITCHING REGULATOR IC
Figure 3.29
A simplified output voltage waveform is shown in Figure 3.30 for a PBM buck converter. Note that the comparator hysteresis voltage multiplied by the reciprocal of the attenuation factor primarily determines the peak-to-peak output voltage ripple (typically between 50 and 100mV). It should be noted that the actual output voltage ripple waveform can look quite different from that shown in Figure 3.30 depending on the design and whether the converter is a buck or boost.
A practical switching regulator IC using the PBM approach is the ADP3000, which has a fixed switching frequency of 400kHz and a fixed duty cycle of 80%. This device is a versatile step-up/step-down converter. It can deliver an output current of 100mA in a 5V to 3V step-down configuration and 180mA in a 2V to 3.3V step-up configuration. Input supply voltage can range between 2V and 12V in the boost mode, and up to 30V in the buck mode. It should be noted that when the oscillator is turned off, the internal switch is opened so that the inductor current does not continue to increase.
3.32
SWITCHING REGULATORS
REPRESENTATIVE OUTPUT VOLTAGE WAVEFORM FOR GATED OSCILLATOR CONTROLLED (PBM) BUCK REGULATORS
VOUT
OSC. ON
R 2 VOUT = VREF 1 + R1
OSC. OFF
R 2 Ripple V hysteresis 1 + R1
0
Figure 3.30 In the gated-oscillator method, the comparator hysteresis serves to stabilize the feedback loop making the designs relatively simple. The disadvantage, of course, is that the peak-to-peak output voltage ripple can never be less than the comparator hysteresis multiplied by the reciprocal of the attenuation factor: R2 Output Ripple Vhysteresis . R1 Because the gated-oscillator (PBM) controlled switching regulator operates with a fixed duty cycle, output regulation is achieved by changing the number of "skipped pulses" as a function of load current and voltage. From this perspective, PBM controlled switchers tend to operate in the "discontinuous" mode under light load conditions. Also, the maximum average duty cycle is limited by the built-in duty cycle of the oscillator. Once the required duty cycle exceeds that limit, no pulse skipping occurs, and the device will lose regulation. One disadvantage of the PBM switching regulator is that the frequency spectrum of the output ripple is "fuzzy" because of the burst-mode of operation. Frequency components may fall into the audio band, so proper filtering of the output of such a regulator is mandatory. Selection of the inductor value is also more critical in PBM regulators. Because the regulation is accomplished with a burst of fixed duty cycle pulses (i.e., higher than needed on average) followed by an extended off time, the energy stored in the inductor during the burst of pulses must be sufficient to supply the required energy to the load. If the inductor value is too large, the regulator may never start up, or may have poor transient response and inadequate line and load regulation. On the other hand, if the inductor value is too small, the inductor may saturate during the charging time, or the peak inductor current may exceed the maximum rated switch 3.33
SWITCHING REGULATORS current. However, devices such as the ADP3000 incorporate on-chip overcurrent protection for the switch. An additional feature allows the maximum peak switch current to be set with an external resistor, thereby preventing inductor saturation. Techniques for selecting the proper inductor value will be discussed in a following section.
DIODE AND SWITCH CONSIDERATIONS
So far, we have based our discussions around an ideal lossless switching regulator having ideal circuit elements. In practice, the diode, switch, and inductor all dissipate power which leads to less than 100% efficiency. Figure 3.31 shows typical buck and boost converters, where the switch is part of the IC. The process is bipolar, and this type of transistor is used as the switching element. The ADP3000 and its relatives (ADP1108, ADP1109, ADP1110, ADP1111, ADP1073, ADP1173) use this type of internal switch.
NPN SWITCHES IN IC REGULATORS ADP1108/1109/1110/1111/1073/1173
VIN + BASE DRIVE iD iSW L C LOAD VOUT BUCK ON-VOLTAGE 1.5V @ 650mA
L VIN + BASE DRIVE
iD VOUT BOOST iSW C LOAD ON-VOLTAGE 1V @ 1A
Figure 3.31 The diode is external to the IC and must be chosen carefully. Current flows through the diode during the off-time of the switching cycle. This translates into an average current which causes power dissipation because of the diode forward voltage drop. The power dissipation can be minimized by selecting a Schottky diode with a low forward drop (0.5V),such as the 1N5818-type. It is also important that the diode capacitance and recovery time be low to prevent additional power loss due to charging current, and this is also afforded by the Schottky diode. Power dissipation can be approximated by multiplying the average diode current by the forward voltage drop.
3.34
SWITCHING REGULATORS The drop across the NPN switch also contributes to internal power dissipation. The power (neglecting switching losses) is equal to the average switch current multiplied by the collector-emitter on-state voltage. In the case of the ADP3000 series, it is 1.5V at the maximum rated switch current of 650mA (when operating in the buck mode). In the boost mode, the NPN switch can be driven into saturation, so the on-state voltage is reduced, and thus, so is the power dissipation. Note that in the case of the ADP3000, the saturation voltage is about 1V at the maximum rated switch current of 1A. In examining the two configurations, it would be logical to use a PNP switching transistor in the buck converter and an NPN transistor in the boost converter in order to minimize switch voltage drop. However, the PNP transistors available on processes which are suitable for IC switching regulators generally have poor performance, so the NPN transistor must be used for both topologies. In addition to lowering efficiency by their power dissipation, the switching transistors and the diode also affect the relationship between the input and output voltage. The equations previously developed assumed zero switch and diode voltage drops. Rather than re-deriving all the equations to account for these drops, we will examine their effects on the inductor current for a simple buck and boost converter operating in the continuous mode as shown in Figure 3.32.
EFFECTS OF SWITCH AND DIODE VOLTAGE ON INDUCTOR CURRENT EQUATIONS
VIN + VSW VD L C BUCK ton VIN + L VSW VD VOUT LOAD INDUCTOR CURRENT toff VOUT
VIN - VOUT - VSW L VOUT + VD t on = t off L
C
LOAD
- VIN + VD V VIN - VSW t off t on = OUT L L
BOOST
Figure 3.32 In the buck converter, the voltage applied to the inductor when the switch is on is equal to VIN VOUT VSW, where VSW is the approximate average voltage drop across the switch. When the switch is off, the inductor current is discharged into a 3.35
SWITCHING REGULATORS voltage equal to VOUT + VD, where VD is the approximate average forward drop across the diode. The basic inductor equation used to derive the relationship between the input and output voltage becomes: VIN - VOUT - VSW VOUT + VD t on = t off . L L In the actual regulator circuit, negative feedback will force the duty cycle to maintain the correct output voltage, but the duty cycle will also be affected by the switch and the diode drops to a lesser degree. When the switch is on in a boost converter, the voltage applied to the inductor is equal to VIN VSW. When the switch is off, the inductor current discharges into a voltage equal to VOUT VIN + VD. The basic inductor current equation becomes: VIN - VSW VOUT - VIN + VD t on = t off . L L From the above equations, the basic relationships between input voltage, output voltage, duty cycle, switch, and diode drops can be derived for the buck and boost converters. The ADP3000 is a switching regulator that uses the NPN-type switch just discussed. A block diagram is shown in Figure 3.33 and key specifications are given in Figure 3.34.
ADP3000 SWITCHING REGULATOR BLOCK DIAGRAM
SET
VIN
A1 GAIN BLOCK/ ERROR AMP
A0
ILIM 1.245V REFERENCE COMPARATOR 400kHz OSCILLATOR R1 R2 DRIVER SW2 SW1
GND
SENSE
Figure 3.33
3.36
SWITCHING REGULATORS
ADP3000 SWITCHING REGULATOR KEY SPECIFICATIONS
n Input Voltages from 2V to 12V (Step-Up), 2V to 30V (Step-Down) n Fixed 3.3V, 5V, 12V and Adjustable Output Voltage n Step-Up or Step-Down Mode n PBM (Gated Oscillator) Control Simplifies Design n 50mV Typical Output Ripple Voltage (5V Output) n 400kHz Switching Frequency Allows Low Value Inductors n 80% Duty Cycle n 500µA Quiescent Current n Output Drive Capability: 100mA @ 3V from 5V Input in Step-Down Mode 180mA @ 3.3V from 2V Input in Step-Up Mode n 8-Pin DIP or SOIC Package Figure 3.34
The device uses the gated oscillator, or pulse burst modulation (PBM) feedback control scheme. The internal oscillator operates at a frequency of 400kHz allowing the use of small value inductors and capacitors. The internal resistors, R1 and R2, set the output voltage to 3.3V, 5V, or 12V, depending upon the option selected. A completely adjustable version is also available where the comparator input is brought out directly to the "SENSE" pin, and the user provides the external divider resistors. Total quiescent current is only 500µA. The uncommitted gain block, A1, can be used as a low-battery detector or to reduce output hysteretic ripple limits by adding gain in the feedback loop. A current-limit pin, ILIM, allows switch current to be limited with an external resistor. Limiting the switch current on a cycle by cycle basis allows the use of small inductors with low saturation current. It also allows physically small tantalum capacitors with a typical ESR of 0.1 to achieve an output ripple voltage as low as 40 to 80mV, as well as low input ripple current. A typical ADP3000 boost application circuit is shown in Figure 3.35. The input voltage can range from +2V to +3.2V. The output is +5V and supplies a load current of 100mA. Typical efficiency for the circuit is 80%. All components are available in surface mount. The ADP3000 can also be used in the buck configuration as shown in Figure 3.36. The input voltage to the regulator is between 5V and 6V, and the output is 3V at 100mA. Note that in this case, the adjustable version of the ADP3000 is used. The external divider resistors, R1 and R2, are chosen to set the nominal output voltage to 3V. All components are ava