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LT1074/LT1076 Step-Down Switching Regulator
FEATURES
s s s s s s s s
5A On-Board Switch (LT1074) 100kHz Switching Frequency Greatly Improved Dynamic Behavior Available in Low Cost 5 and 7-Lead Packages Only 8.5mA Quiescent Current Programmable Current Limit Operates Up to 60V Input Micropower Shutdown Mode
tions allow this device to be used as a positive-to-negative converter, a negative boost converter, and as a flyback converter. The switch output is specified to swing 40V below ground, allowing the LT1074 to drive a tappedinductor in the buck mode with output currents up to 10A. The LT1074 uses a true analog multiplier in the feedback loop. This makes the device respond nearly instantaneously to input voltage fluctuations and makes loop gain independent of input voltage. As a result, dynamic behavior of the regulator is significantly improved over previous designs. On-chip pulse by pulse current limiting makes the LT1074 nearly bust-proof for output overloads or shorts. The input voltage range as a buck converter is 8V to 60V, but a selfboot feature allows input voltages as low as 5V in the inverting and boost configurations. The LT1074 is available in low cost TO-220 or TO-3 packages with frequency pre-set at 100kHz and current limit at 6.5A (LT1076 = 2.6A). A 7-pin TO-220 package is also available which allows current limit to be adjusted down to zero. In addition, full micropower shutdown can be programmed. See Application Note 44 for design details. A fixed 5V output, 2A version is also available. See LT1076-5.
APPLICATI
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Buck Converter with Output Voltage Range of 2.5V to 50V Tapped-Inductor Buck Converter with 10A Output at 5V Positive-to-Negative Converter Negative Boost Converter Multiple Output Buck Converter
DESCRIPTIO
The LT1074 is a 5A (LT1076 is rated at 2A) monolithic bipolar switching regulator which requires only a few external parts for normal operation. The power switch, all oscillator and control circuitry, and all current limit components, are included on the chip. The topology is a classic positive "buck" configuration but several design innova-
TYPICAL APPLICATI
Basic Positive Buck Converter
L1** 50 µH (LT1074) 100 µH (LT1076)
10V TO 40V
LT1074 GND VC FB R3 2.7k
C3 200 µF
MBR745*
R1 2.8k 1% R2 2.21k 1% +
+
C2 0.01µ F
C1 500 µF 25V
*USE MBR340 FOR LT1076 **COILTRONICS #50-2-52 (LT1074) #100-1-52 (LT1076) PULSE ENGINEERING, INC. #PE-92114 (LT1074) #PE-92102 (LT1076) HURRICANE #HL-AK147QQ (LT1074) #HL-AG210LL (LT1076) RIPPLE CURRENT RATING I OUT / 2
EFFICIENCY (%)
VIN
VSW
UO
U
UO
S
Buck Converter Efficiency
LT1074 100
5V 5A
90 80
VOUT = 12V, VIN = 20V
VOUT = 5V, VIN = 15V 70 60 50 0 1 2 3 4 5 6 OUTPUT LOAD CURRENT (A)
LT1074 · TPC27
L = 50 µ H TYPE 52 CORE DIODE = MBR735
LT1074 · TA01
1
LT1074/LT1076 ABSOLUTE AXI U RATI GS
ILIM Pin Voltage (Forced) ............................................ 5.5V Maximum Operating Ambient Temperature Range LT1074C/76C, LT1074HVC/76HVC ............ 0°C to 70°C LT1074I/76I, LT1074HVI/76HVI ............. 40°C to 85°C LT1074M/76M, LT1074HVM/76HVM ... 55°C to 125°C Maximum Operating Junction Temperature Range LT1074C/76C, LT1074HVC/76HVC .......... 0°C to 125°C LT1074I/76I, LT1074HVI/76HVI ........... 40°C to 125°C LT1074M/76M, LT1074HVM/76HVM ... 55°C to 150°C Maximum Storage Temperature ................ 65°C to 150°C Lead Temperature (Soldering, 10 sec) ..................... 300°C Input Voltage LT1074/ LT1076 .................................................. 45V LT1074HV/76HV.................................................. 64V Switch Voltage with Respect to Input Voltage LT1074/ 76 .......................................................... 64V LT1074HV/76HV.................................................. 75V Switch Voltage with Respect to Ground Pin (VSW Negative) LT1074/76 (Note 6) ............................................. 35V LT1074HV/76HV (Note 6).................................... 45V Feedback Pin Voltage ..................................... 2V, +10V Shutdown Pin Voltage (Not to Exceed VIN) .............. 40V
PACKAGE/ORDER I FOR ATIO
FRONT VIEW 5 4 3 2 1 VIN VSW GND VC FB/SENSE
ORDER PART NUMBER
BOTTOM VIEW
LT1076CQ
Q PACKAGE 5-LEAD PLASTIC DD LT1076: JC = 4°C/W, JA = 30°C/W*
FRONT VIEW 7 6 5 4 3 2 1 R PACKAGE 7-LEAD PLASTIC DD LT1076: JC = 4°C/W, JA = 30°C/W*
FRONT VIEW
6 4 2 7 5 3 1
SHDN VC FB/SENSE GND ILIM VSW VIN
LT1076CR LT1076HVCR
FB GND ILIM VSW VIN
VC
SHUTDOWN
Y PACKAGE, 7-LEAD TO-220 LT1074: JC = 2.5°C/W, JA = 50°C/W LT1076: JC = 4°C/W, JA = 50°C/W
LT1074CY LT1074HVCY LT1074IY LT1074HVIY LT1076CY LT1076HVCY
* Assumes package is soldered to 0.5 IN2 of 1 oz. copper over internal ground plane or over back side plane.
ELECTRICAL CHARACTERISTICS
PARAMETER Switch "On" Voltage (Note 1) CONDITIONS LT1074
Tj = 25°C, VIN = 25V, unless otherwise noted.
MIN TYP MAX 1.85 2.1 2.3 2.5 q q 1.2 1.7 UNITS V V V V V V
ISW = 1A, Tj 0°C ISW = 1A, Tj < 0°C ISW = 5A, Tj 0°C ISW = 5A, Tj < 0°C ISW = 0.5A ISW = 2A
LT1076
2
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ORDER PART NUMBER
VC 1 4 FB 2 3 VIN CASE IS GND
VSW K PACKAGE, 4-LEAD TO-3 METAL CAN LT1074: JC = 2.5°C/W, JA = 35°C/W LT1076: JC = 4°C/W, JA = 35°C/W
LT1074CK LT1074HVCK LT1074MK LT1074HVMK LT1076CK LT1076HVCK LT1076MK LT1076HVMK LT1074CT LT1074HVCT LT1074IT LT1074HVIT LT1076CT LT1076HVCT LT1076IT
FRONT VIEW
5 4 3 2 1
VSW GND VC FB
VIN
T PACKAGE, 5-LEAD T0-220 LEADS ARE FORMED STANDARD FOR STRAIGHT LEADS, ORDER FLOW 06 LT1074: JC = 2.5°C/W, JA = 50°C/W LT1076: JC = 4°C/W, JA = 50°C/W
LT1074/LT1076
ELECTRICAL CHARACTERISTICS
PARAMETER Switch "Off" Leakage CONDITIONS LT1074 LT1076 Supply Current (Note 2) VIN 25V, VSW = 0 VIN = VMAX, VSW = 0 (Note 7) VIN = 25V, VSW = 0 VIN = VMAX, VSW = 0 (Note 7) q q q q q q 5.5 8.5 9 140 7.3 3.5 6.5 4.5 3 2.6 1.8 1.2 90 100 110 120 125 0.1
Tj = 25°C, VIN = 25V, unless otherwise noted.
MIN TYP 5 10 MAX 300 500 150 250 11 12 300 8 4.8 8.5 UNITS µA µA µA µA mA mA µA V V A A A A A A % kHz kHz kHz kHz %/V V/V 8000 225 1.6 2 2.265 ± 1.5 ± 2.5 0.02 µmho µA mA µA V % % %/V V mV/°C V 20 50 2.7 0.5 2.5 4.0 µA µA V V °C/W °C/W
VFB = 2.5V, VIN 40V 40V < VIN < 60V VSHUT = 0.1V (Device Shutdown) (Note 8) Normal Mode Startup Mode (Note 3) LT1074 ILIM Open RLIM = 10k (Note 5) RLIM = 7k (Note 5) ILIM Open RLIM = 10k (Note 5) RLIM = 7k (Note 5)
Minimum Supply Voltage Switch Current Limit (Note 4)
LT1076
q
2
3.2
Maximum Duty Cycle Switching Frequency Tj 125°C Tj > 125°C VFB = 0V through 2k (Note 4) 8V VIN VMAX (Note 7) 1V VC 4V
q q q q
85 90 85 85
20 0.03 2000 3700 5000 140 1 0.5 2.155 2.21 ± 0.5 ±1 0.005 1.5 4 24
Switching Frequency Line Regulation Error Amplifier Voltage Gain (Note 6) Error Amplifier Transconductance Error Amplifier Source and Sink Current Feedback Pin Bias Current Reference Voltage Reference Voltage Tolerance
Source (VFB = 2V) Sink (VFB = 2.5V) VFB = VREF VC = 2V VREF (Nominal) = 2.21V All Conditions of Input Voltage, Output Voltage, Temperature and Load Current 8V VIN VMAX (Note 7) Over Temperature q q q q q
100 0.7
Reference Voltage Line Regulation VC Voltage at 0% Duty Cycle Multiplier Reference Voltage Shutdown Pin Current Shutdown Thresholds Thermal Resistance Junction to Case
VSH = 5V VSH VTHRESHOLD (2.5V) Switch Duty Cycle = 0 Fully Shut Down LT1074 LT1076
q q q q
5 2.2 0.1
10 2.45 0.3
The q denotes the specifications which apply over the full operating temperature range. Note 1: To calculate maximum switch "on" voltage at currents between low and high conditions, a linear interpolation may be used. Note 2: A feedback pin voltage (VFB) of 2.5V forces the VC pin to its low clamp level and the switch duty cycle to zero. This approximates the zero load condition where duty cycle approaches zero. Note 3: Total voltage from VIN pin to ground pin must be 8V after startup for proper regulation.
Note 4: Switch frequency is internally scaled down when the feedback pin voltage is less than 1.3V to avoid extremely short switch on times. During testing, VFB is adjusted to give a minimum switch on time of 1µs. Note 5: ILIM RLIM 1k R 1k (LT1074), ILIM LIM (LT1076). 2k 5.5k
Note 6: Switch to input voltage limitation must also be observed. Note 7: VMAX = 40V for the LT1074/76 and 60V for the LT1074HV/76HV. Note 8: Does not include switch leakage.
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LT1074/LT1076
BLOCK DIAGRA
INPUT SUPPLY
10 µ A
0.3V + µ-POWER SHUTDOWN 6V REGULATOR AND BIAS 6V TO ALL CIRCUITRY CURRENT LIMIT COMP C2 500
2.35V +
SHUTDOWN*
+
2.21V
A1 ERROR AMP
FB
*AVAILABLE ON PACKAGES WITH PIN COUNTS GREATER THAN 5.
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LT1074 320 µ A CURRENT LIMIT SHUTDOWN + 250 0.04 I LIM* 4.5V 10k FREQ SHIFT 100kHz OSCILLATOR SYNC 3V(p-p) VIN + Z ANALOG X MULTIPLIER XY Z Y C1 PULSE WIDTH COMPARATOR 400 15 R R/S Q S LATCH R G1 SWITCH OUTPUT (VSW ) LT1076 VC 24V (EQUIVALENT) 0.1 100 SWITCH OUTPUT (VSW )
LT1074 · BD01
LT1074/LT1076 BLOCK DIAGRA DESCRIPTIO
A switch cycle in the LT1074 is initiated by the oscillator setting the R/S latch. The pulse that sets the latch also locks out the switch via gate G1. The effective width of this pulse is approximately 700ns, which sets the maximum switch duty cycle to approximately 93% at 100kHz switching frequency. The switch is turned off by comparator C1, which resets the latch. C1 has a sawtooth waveform as one input and the output of an analog multiplier as the other input. The multiplier output is the product of an internal reference voltage, and the output of the error amplifier, A1, divided by the regulator input voltage. In standard buck regulators, this means that the output voltage of A1 required to keep a constant regulated output is independent of regulator input voltage. This greatly improves line transient response, and makes loop gain independent of input voltage. The error amplifier is a transconductance type with a GM at null of approximately 5000µmho. Slew current going positive is 140µA, while negative slew current is about 1.1mA. This asymmetry helps prevent overshoot on start-up. Overall loop frequency compensation is accomplished with a series RC network from VC to ground. Switch current is continuously monitored by C2, which resets the R/S latch to turn the switch off if an overcurrent condition occurs. The time required for detection and switch turn off is approximately 600ns. So minimum switch "on" time in current limit is 600ns. Under dead shorted output conditions, switch duty cycle may have to be as low as 2% to maintain control of output current. This would require switch on time of 200ns at 100kHz switching frequency, so frequency is reduced at very low output
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voltages by feeding the FB signal into the oscillator and creating a linear frequency downshift when the FB signal drops below 1.3V. Current trip level is set by the voltage on the ILIM pin which is driven by an internal 320µA current source. When this pin is left open, it self-clamps at about 4.5V and sets current limit at 6.5A for the LT1074 and 2.6A for the LT1076. In the 7-pin package an external resistor can be connected from the ILIM pin to ground to set a lower current limit. A capacitor in parallel with this resistor will soft start the current limit. A slight offset in C2 guarantees that when the ILIM pin is pulled to within 200mV of ground, C2 output will stay high and force switch duty cycle to zero. The "Shutdown" pin is used to force switch duty cycle to zero by pulling the ILIM pin low, or to completely shut down the regulator. Threshold for the former is approximately 2.35V, and for complete shutdown, approximately 0.3V. Total supply current in shutdown is about 150µA. A 10µA pull-up current forces the shutdown pin high when left open. A capacitor can be used to generate delayed startup. A resistor divider will program "undervoltage lockout" if the divider voltage is set at 2.35V when the input is at the desired trip point. The switch used in the LT1074 is a Darlington NPN (single NPN for LT1076) driven by a saturated PNP. Special patented circuitry is used to drive the PNP on and off very quickly even from the saturation state. This particular switch arrangement has no "isolation tubs" connected to the switch output, which can therefore swing to 40V below ground.
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LT1074/LT1076
TYPICAL PERFOR A CE CHARACTERISTICS
VC Pin Characteristics
200 150 100 VFB ADJUSTED FOR IC = 0 AT VC = 2V 2.0 1.5 1.0
CURRENT (mA)
CURRENT (mA)
50 0 50 SLOPE 400k VFB 2V
0.5 0 0.5 1.0 1.5 2.0
CURRENT (µA)
100 150 200 0 1 2
3
4
5
6
VOLTAGE (V)
LT1074 · TPC01
Shutdown Pin Characteristics
40 30 20
CURRENT ( µ A) CURRENT ( µ A)
10 0 10 20 30 40 0 10 20
15 20 25 30
CURRENT ( µ A)
VIN = 50V THIS POINT MOVES WITH VIN
DETAILS OF THIS AREA SHOWN IN OTHER GRAPH 30 40 50 60 70 80
VOLTAGE (V)
LT1074 · TPC04
INPUT CURRENT (mA)
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7 8 9
VC Pin Characteristics
Feedback Pin Characteristics
500 400 300
VFB 2.5V
200 100 0 100 200 300 400 500
START OF FREQUENCY SHIFTING
0
1
2
3
4
5
6
7
8
9
0
1
2
3
4
5
6
7
8
9
10
VOLTAGE (V)
LT1074 · TPC02
VOLTAGE (V)
LT1074 · TPC03
Shutdown Pin Characteristics
0 5 10 Tj = 25°C CURRENT FLOWS OUT OF SHUTDOWN PIN
ILIM Pin Characteristics
100 50 0 50 Tj = 25°C
SHUTDOWN THRESHOLD
100 150 200 250 300
35 40 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 VOLTAGE (V)
LT1074 · TPC05
350 400 2 1 0 1 2 3 4 5 6 7 8
VOLTAGE (V)
LT1074 · TPC06
Supply Current
20 18 16 14 12 10 8 6 4 2 0 0 10 20 30 40 50 60 INPUT VOLTAGE (V)
LT1074 · TPC11
DEVICE NOT SWITCHING VC = 1V
LT1074/LT1076
TYPICAL PERFOR A CE CHARACTERISTICS
Supply Current (Shutdown)
300 250
2.25 2.24 2.23 VOLTAGE (V)
INPUT CURRENT ( µ A)
200 150 100 50 0 0 10 20 30 40 50 60 INPUT VOLTAGE (V)
LT1074 · TPC13
2.22 2.21 2.20 2.19
"ON" VOLTAGE (V)
Reference Shift with Ripple Voltage
20
CHANGE IN REFERENCE VOLTAGE (mV)
10 0 10 20 30 40 50 60 70 80 0 20 40 60 80 100 120 140 160 180 200
LT1074 · TPC16
TRANSCONDUCTANCE ( µ mho)
TRI WAVE SQUARE WAVE
FREQUENCY (kHz)
PEAK-TO-PEAK RIPPLE AT FB PIN (mV)
Feedback Pin Frequency Shift
160 140
8 7
SWITCHING FREQUENCY (kHz)
OUTPUT CURRENT LIMIT (A)
120 100 80 150°C 60 40 20 0 0 0.5 1.0 1.5 2.0 2.5 3.0 FEEDBACK PIN VOLTAGE (V)
LT1074 · TPC19
55°C 25°C
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Reference Voltage vs Temperature
3.0
Switch "On" Voltage
Tj = 25° C 2.5
2.0
LT1074
1.5 LT1076 1.0
2.18 2.17 50 25 0 25 50 75 100 125 150
LT1074 · TPC14
0.5
0
1
2
3
4
5
6
JUNCTION TEMPERATURE (°C)
SWITCH CURRENT (A)
LT1074 · TPC28
8k 7k 6k 5k 4k 3k 2k 1k 0
Error Amplifier Phase and GM
Switching Frequency vs Temperature
200 150 120 115 110 105 100 95 90 85 80 50 25 0 25 50 75 100 125 150
LT1074 · TPC18
100 50
PHASE ( °)
GM
0
50
100 150 1k 10k 100k FREQUENCY (Hz)
LT1074 · TPC17
1M
200 10M
JUNCTION TEMPERATURE ( ° C)
Current Limit vs Temperature*
I LIM PIN OPEN
6 5 4 3 2 1 *MULTIPLY CURRENTS BY 0.4 FOR LT1076 0 50 25 0 25 50 75 100 125 150 JUNCTION TEMPERATURE (° C)
LT1074 · TPC22
R LIM = 10k
RLIM = 5k
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LT1074/LT1076
PI DESCRIPTIO S
VIN PIN The VIN pin is both the supply voltage for internal control circuitry and one end of the high current switch. It is important, especially at low input voltages, that this pin be bypassed with a low ESR, and low inductance capacitor to prevent transient steps or spikes from causing erratic operation. At full switch current of 5A, the switching transients at the regulator input can get very large as shown in Figure 1. Place the input capacitor very close to the regulator and connect it with wide traces to avoid extra inductance. Use radial lead capacitors. VOUT =
( dI)(L P) dt
STEP =
( I SW )(ESR)
Figure 1. Input Capacitor Ripple
LP = Total inductance in input bypass connections and capacitor. "Spike" height (dI/dt · LP) is approximately 2V per inch of lead length for LT1074 and 0.8V per inch for LT1076. "Step" for ESR = 0.05 and ISW = 5A is 0.25V. "Ramp" for C = 200µF, TON = 5µs, and ISW = 5A, is 0.12V. Input current on the VIN Pin in shutdown mode is the sum of actual supply current (140µA, with a maximum of 300µA), and switch leakage current. Consult factory for special testing if shutdown mode input current is critical. GROUND PIN It might seem unusual to describe a ground pin, but in the case of regulators, the ground pin must be connected properly to ensure good load regulation. The internal reference voltage is referenced to the ground pin; so any error in ground pin voltage will be multiplied at the output;
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(VGND ) (VOUT )
2 .21
To ensure good load regulation, the ground pin must be connected directly to the proper output node, so that no high currents flow in this path. The output divider resistor should also be connected to this low current connection line as shown in Figure 2.
LT1074 GND FB
R2
RAMP =
( I SW )(TON )
C
LT1074 · PD01
HIGH CURRENT RETURN PATH
NEGATIVE OUTPUT NODE WHERE LOAD REGULATION WILL BE MEASURED
LT1074 · PD02
Figure 2. Proper Ground Pin Connection
FEEDBACK PIN The feedback pin is the inverting input of an error amplifier which controls the regulator output by adjusting duty cycle. The non-inverting input is internally connected to a trimmed 2.21V reference. Input bias current is typically 0.5µA when the error amplifier is balanced (IOUT = 0). The error amplifier has asymmetrical GM for large input signals to reduce startup overshoot. This makes the amplifier more sensitive to large ripple voltages at the feedback pin. 100mVp-p ripple at the feedback pin will create a 14mV offset in the amplifier, equivalent to a 0.7% output voltage shift. To avoid output errors, output ripple (P-P) should be less than 4% of DC output voltage at the point where the output divider is connected. See the "Error Amplifier" section for more details. Frequency Shifting at the Feedback Pin The error amplifier feedback pin (FB) is used to downshift the oscillator frequency when the regulator output voltage
LT1074/LT1076
PI DESCRIPTIO S
is low. This is done to guarantee that output short circuit current is well controlled even when switch duty cycle must be extremely low. Theoretical switch "on" time for a buck converter in continuous mode is; SHUTDOWN PIN The shutdown pin is used for undervoltage lockout, micropower shutdown, soft start, delayed start, or as a general purpose on/off control of the regulator output. It controls switching action by pulling the ILIM pin low, which forces the switch to a continuous "off" state. Full micropower shutdown is initiated when the shutdown pin drops below 0.3V. The V/I characteristics of the shutdown pin are shown in Figure 4. For voltages between 2.5V and VIN, a current of 10µA flows out of the shutdown pin. This current increases to 25µA as the shutdown pin moves through the 2.35V threshold. The current increases further to 30µA at the 0.3V threshold, then drops to 15µA as the shutdown voltage falls below 0.3V. The 10µA current source is included to pull the shutdown pin to its high or default state when left open. It also provides a convenient pullup for delayed start applications with a capacitor on the shutdown pin. When activated, the typical collector current of Q1 in Figure 5, is 2mA. A soft start capacitor on the ILIM pin will delay regulator shutdown in response to C1, by (5V)(CLIM)/2mA. Soft start after full micropower shutdown is ensured by coupling C2 to Q1.
0
LT1074 · PD03
V + VD t ON = OUT VIN · f
VD = Catch diode forward voltage ( 0.5V) f = Switching frequency At f = 100kHz, tON must drop to 0.2µs when VIN = 25V and the output is shorted (VOUT = 0V). In current limit, the LT1074 can reduce tON to a minimum value of 0.6µs, much too long to control current correctly for V OUT = 0. To correct this problem, switching frequency is lowered from 100kHz to 20kHz as the FB pin drops from 1.3V to 0.5V. This is accomplished by the circuitry shown in Figure 3. TO
OSCILLATOR VOUT +2V + ERROR AMPLIFIER 2.21V Q1 R3 3k R1
VC
Figure 3. Frequency Shifting
Q1 is off when the output is regulating (VFB = 2.21V). As the output is pulled down by an overload, VFB will eventually reach 1.3V, turning on Q1. As the output continues to drop, Q1 current increases proportionately and lowers the frequency of the oscillator. Frequency shifting starts when the output is 60% of normal value, and is down to its minimum value of 20kHz when the output is 20% of normal value. The rate at which frequency is shifted is determined by both the internal 3k resistor R3 and the external divider resistors. For this reason, R2 should not be increased to more than 4k, if the LT1074 will be subjected to the simultaneous conditions of high input voltage and output short circuit.
CURRENT ( µ A)
U
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EXTERNAL DIVIDER FB R2 2.21k
5 10 15 20 25 30 35 40 0
Tj = 25°C CURRENT FLOWS OUT OF SHUTDOWN PIN
SHUTDOWN THRESHOLD
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
VOLTAGE (V)
LT1074 · TPC05
Figure 4. Shutdown Pin Characteristics
9
LT1074/LT1076 PI DESCRIPTIO S
V IN 10 µ A 300 µ A
SHUTDOWN PIN 2.3V
C1 +
C2 0.3V + TO TOTAL REGULATOR SHUTDOWN
LT1074 · PD07
Figure 5. Shutdown Circuitry
Undervoltage Lockout Undervoltage lockout point is set by R1 and R2 in Figure 6. To avoid errors due to the 10µA shutdown pin current, R2 is usually set at 5k, and R1 is found from:
R1 = R 2
(VTP VSH )
VSH
VTP = Desired undervoltage lockout voltage. VSH = Threshold for lockout on the shutdown pin = 2.45V. If quiescent supply current is critical, R2 may be increased up to 15k, but the denominator in the formula for R2 should replace VSH with VSH (10µA)(R2).
R1 SHUT
LT1074
R2 5k
Figure 6. Undervoltage Lockout
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Hysteresis in undervoltage lockout may be accomplished by connecting a resistor (R3) from the ILIM pin to the shutdown pin as shown in Figure 7. D1 prevents the shutdown divider from altering current limit.
R1 SHUT
EXTERNAL CLIM
ILIM PIN
VIN
Q1
6V
R3
D1* I LIM
LT1074
R2
OPTIONAL CURRENT LIMIT RESISTOR
LT1074 · PD09
*1N4148
Figure 7. Adding Hysteresis
R1 Trip Point = VTP = 2 . 35 V 1 + R 2 If R3 is added, the lower trip point (VIN descending) will be the same. The upper trip point (VUTP) will be; R1 R1 R1 VUTP = VSH 1 + + 0. 8 V R 3 R 2 R 3 If R1 and R2 are chosen, R3 is given by
R3 =
(VSH 0.8 V) (R1)
R1 VUTP VSH 1 + R 2
VIN
Example: An undervoltage lockout is required such that the output will not start until VIN = 20V, but will continue to operate until VIN drops to 15V. Let R2 = 2.32k.
R1 = 2 . 32k R3 =
GND
(
LT1074 · PD08
(
2 . 35 V 2 . 35 0. 8 12 . 5
)
(15V 2.35V) = 12.5k )( )
= 3. 9 k
12 . 5 20 2 . 35 1 + 2 . 32
LT1074/LT1076
PI DESCRIPTIO S
ILIM PIN The ILIM pin is used to reduce current limit below the preset value of 6.5A. The equivalent circuit for this pin is shown in Figure 8.
TO LIMIT CIRCUIT VIN 320 µ A D2 Q1 D1 R1 8K D3 6V I LIM
LT1047 · PD12
Figure 8. ILIM Pin Circuit
LT1074
When ILIM is left open, the voltage at Q1 base clamps at 5V through D2. Internal current limit is determined by the current through Q1. If an external resistor is connected between ILIM and ground, the voltage at Q1 base can be reduced for lower current limit. The resistor will have a voltage across it equal to (320µA) (R), limited to 5V when clamped by D2. Resistance required for a given current limit is RLIM = ILIM (2k) + 1k (LT1074) RLIM = ILIM (5.5k) + 1k (LT1076) As an example, a 3A current limit would require 3A (2k) + 1k = 7k for the LT1074. The accuracy of these formulas is ±25% for 2A ILIM 5A (LT1074) and 0.7A ILIM 1.8A (LT1076), so ILIM should be set at least 25% above the peak switch current required. Foldback current limiting can be easily implemented by adding a resistor from the output to the ILIM pin as shown in Figure 9. This allows full desired current limit (with or without RLIM) when the output is regulating, but reduces current limit under short circuit conditions. A typical value for RFB is 5k, but this may be adjusted up or down to set the amount of foldback. D2 prevents the output voltage
U
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from forcing current back into the ILIM pin. To calculate a value for RFB, first calculate RLIM, then RFB; RFB =
(ISC - 0.44*) (RL ) (R ink) L 0.5 * (RL 1k) ISC
*Change 0.44 to 0.16, and 0.5 to 0.18 for LT1076. Example: ILIM = 4A, ISC = 1.5A, RLIM = (4)(2k) + 1k = 9k
4.3V
R FB =
. (15 - 0.44) (9k) . 0.5 (9k - 1k ) - 15
VOUT
I LIM
FB
R FB R LIM
D2 1N4148
LT1074 · PD13
Figure 9. Foldback Current Limit
ERROR AMPLIFIER The error amplifier in Figure 10 is a single stage design with added inverters to allow the output to swing above and below the common mode input voltage. One side of the amplifier is tied to a trimmed internal reference voltage of 2.21V. The other input is brought out as the FB (feedback) pin. This amplifier has a GM (voltage "in" to current "out") transfer function of 5000µmho. Voltage gain is determined by multiplying GM times the total equivalent output loading, consisting of the output resistance of Q4 and Q6 in parallel with the series RC external frequency compensation network. At DC, the external RC is ignored, and with a parallel output impedance for Q4 and Q6 of 400k, voltage gain is 2000. At frequencies above a few hertz, voltage gain is determined by the external compensation, RC and CC.
11
LT1074/LT1076
PI DESCRIPTIO S
5.8V
AV =
Gm at midfrequencies 2 · f · CC
A V = Gm· RC at high frequencies
Phase shift from the FB pin to the VC pin is 90° at midfrequencies where the external CC is controlling gain, then drops back to 0° (actually 180° since FB is an inverting input) when the reactance of CC is small compared to RC. The low frequency "pole" where the reactance of CC is equal to the output impedance of Q4 and Q6 (rO), is
1 fPOLE = r0 400k 2 · r0 · C
Although fPOLE varies as much as 3:1 due to rO variations, mid-frequency gain is dependent only on GM, which is specified much tighter on the data sheet. The higher frequency "zero" is determined solely by RC and CC.
fZERO = 1 2 · RC · CC
12
U
U
Q4 90 µ A Q3 50 µ A D1 FB 50 µ A D2 Q6 2.21V 140 µ A 300 CC VC
90 µ A
Q1 X1.8
Q2
90 µ A RC
EXTERNAL FREQUENCY COMPENSATION
ALL CURRENTS SHOWN ARE AT NULL CONDITION
LT1074 · PD11
Figure 10. Error Amplifier
The error amplifier has asymmetrical peak output current. Q3 and Q4 current mirrors are unity gain, but the Q6 mirror has a gain of 1.8 at output null and a gain of 8 when the FB pin is high (Q1 current = 0). This results in a maximum positive output current of 140µA and a maximum negative (sink) output current of 1.1mA. The asymmetry is deliberate -- it results in much less regulator output overshoot during rapid start-up or following the release of an output overload. Amplifier offset is kept low by area scaling Q1 and Q2 at 1.8:1. Amplifier swing is limited by the internal 5.8V supply for positive outputs and by D1 and D2 when the output goes low. Low clamp voltage is approximately one diode drop ( 0.7V 2mV/°C). Note that both the FB pin and the VC pin have other internal connections. Refer to the frequency shifting and sychronizing discussions.
LT1074/LT1076
TYPICAL APPLICATI
20V - 35V LT1074HV GND VC FB R3 1k C2 0.2 µ F
VIN
+
* = 1% FILM RESISTORS D1 = MOTOROLA-MBR745 C1 = NICHICON-UPL1C221MRH6 C2 = NICHICON-UPL1A102MRH6 L1 = COILTRONICS-CTX25-5-52
UO
+
S
Tapped-Inductor Buck Converter
L2 5µH 1 D1** R1 2.8k
L1* VIN VSW D2 35V 5W 3
VOUT 5V, 10A
+
D3 1N5819 0.01µF R2 2.21k
C1 4400 µF (2 EA 2200 µF, 16V)
+
C4 390 µF 16V
C3 200 µ F 50V
*PULSE ENGINEERING #PE65282 **MOTOROLA MBR2030CTL IF INPUT VOLTAGE IS BELOW 20V, MAXIMUM OUTPUT CURRENT WILL BE REDUCED. SEE AN44
LT1074 · TA02
Positive-to-Negative Converter with 5V Output
VIN 4.5V to 40V
+
C1 220µF 50V L1 25µH 5A ¦ ¦
VIN
LT1074
VSW
R1** 5.1k R2** 10k
R3* 2.74k
+
VFB D1 MBR745 C3 0.1µF C4** 0.01µF
¦
C2 1000 µ F 10V OPTIONAL FILTER 5µH
GND
VC
R4 1.82k* 5V,1A***
200µF + 10V
¦
LOWER REVERSE VOLTAGE RATING MAY BE USED FOR LOWER INPUT VOLTAGES. LOWER CURRENT RATING IS ALLOWED FOR LOWER OUTPUT CURRENT. SEE AN44. LOWER CURRENT RATING MAY BE USED FOR LOWER OUTPUT CURRENT. SEE AN44. BUT R1 AND R2 MUST BE INCLUDED IN THE CALCULATION FOR OUTPUT VOLTAGE DIVIDER VALUES. FOR HIGHER OUTPUT VOLTAGES, INCREASE R1, R2, AND R3 PROPORTIONATELY. FOR INPUT VOLTAGE > 10V, R1, R2, AND C4 CAN BE ELIMINATED, AND COMPENSATION IS DONE TOTALLY ON THE VC PIN. R3 = VOUT 2.37 (K ) R1 = (R3) (1.86) R2 = (R3) (3.65)
¦¦
** R1, R2, AND C4 ARE USED FOR LOOP FREQUENCY COMPENSATION WITH LOW INPUT VOLTAGE,
* ** MAXIMUM OUTPUT CURRENT OF 1A IS DETERMINED BY MINIMUM INPUT VOLTAGE OF 4.5V. HIGHER MINIMUM INPUT VOLTAGE WILL ALLOW MUCH HIGHER OUTPUT CURRENTS. SEE AN44.
LT1074 · TA03
13
LT1074/LT1076
TYPICAL APPLICATI UO
+
C3 0.01µF
S
Negative Boost Converter
VIN
100pF FB
R1 12.7k
LT1074 GND 200µF 15V V VC SW R2 2.21k
+
+ C2
1nF R3 750
L1 25µH
D1*
C1 1000 µ F 25V
VIN 5 TO 15V *MBR735 ** IOUT (MAX) = 1A-3A DEPENDING ON INPUT VOLTAGE. SEE AN44
VOUT** 15V
+
5µH
100µF
OPTIONAL OUTPUT FILTER
LT1074 · TA04
PACKAGE DESCRIPTIO
Q Package, 5-Lead PLASTIC DD
0.060 (1.524)
0.401 ± 0.015 (10.185 ± 0.381) 0.175 ± 0.008 (4.445 ± 0.203) 15° TYP
(
+0.012 0.331 0.020
+0.305 8.407 0.508
)
0.067 ± 0.010 (1.702 ± 0.254) 0.032 ± 0.008 (0.813 ± 0.203)
0.059 (1.499) TYP
(
+0.012 0.143 0.020
+0.305 3.632 0.508
)
0.022 ± 0.005 (0.559 ± 0.127)
14
U
Dimensions in inches (milimeters) unless otherwise noted.
R Package, 7-Lead PLASTIC DD
0.060 (1.524)
0.401 ± 0.015 (10.185 ± 0.381)
0.050 ± 0.008 (1.270 ± 0.203) +0.008 0.004 0.004
+0.012 0.331 0.020
0.175 ± 0.008 (4.445 ± 0.203)
15° TYP
0.050 ± 0.008 (1.270 ± 0.203) +0.008 0.004 0.004
(
+0.203 0.102 0.102
)
(
+0.305 8.407 0.508
)
0.050 ± 0.010 (1.270 ± 0.254) 0.030 ± 0.008 (0.762 ± 0.203)
0.059 (1.499) TYP
(
+0.203 0.102 0.102
)
0.105 ± 0.008 (2.667 ± 0.203)
0.105 ± 0.008 (2.667 ± 0.203)
0.050 ± 0.012 (1.270 ± 0.305)
DD5 0693
(
+0.012 0.143 0.020
+0.305 3.632 0.508
)
0.022 ± 0.005 (0.559 ± 0.127)
0.050 ± 0.012 (1.270 ± 0.305)
DD7 0693
LT1074/LT1076
PACKAGE DESCRIPTIO
0.320 0.350 (8.13 8.89)
0.420 0.480 (10.67 12.19)
0.380 0.420 (9.652 10.668) 0.139 0.153 (3.531 3.886) DIA
0.560 0.650 (14.224 16.510) 0.866 0.913 (21.996 23.190)
0.057 0.077 (1.448 1.956)
0.390 0.410 (9.91 10.41)
0.103 0.113 (2.62 2.87)
0.026 0.036 (0.66 0.91) 0.045 0.055 (1.14 1.40)
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
U
Dimensions in inches (milimeters) unless otherwise noted. K Package, 4-Lead TO-3 Metal Can
1.177 1.197 (29.90 30.40) 0.060 0.135 (1.524 3.429) 0.655 0.675 (16.64 19.05) 0.151 0.161 (3.84 4.09) DIA 2 PLC 0.167 0.177 (4.24 4.49) R TYP 72° 18° 0.495 0.525 (12.57 13.34) R
K4 0592
0.760 0.775 (19.30 19.69)
0.470 TP P.C.D.
0.038 0.043 (0.965 1.09)
T Package, 5-Lead TO-220
0.079 0.135 (2.007 3.429) 0.169 0.185 (4.293 4.699) 0.035 0.055 (0.889 1.397)
0.460 0.500 (11.68 12.70)
0.620 ± 0.020 (15.75 ± 0.508) 0.700 0.728 (17.780 18.491)
0.970 1.050 (24.64 26.67)
0.028 0.035 (0.711 0.889)
0.015 0.025 (0.381 0.635) 0.210 0.240 (5.334 6.096) 0.304 0.380 (7.722 9.652)
0.055 0.090 (1.397 2.286) 0.079 0.115 (2.007 2.921) 0.149 0.230 (3.785 5.842)
T5 (FORMED) 0993
Y Package, 7-Lead Molded TO-220
0.147 0.155 (3.73 3.94) DIA 0.169 0.185 (4.29 4.70) 0.045 0.055 (1.14 1.40)
0.235 0.258 (5.97 6.55) 0.560 0.590 (14.22 14.99) 0.620 (15.75) TYP 0.700 0.728 (17.78 18.49)
0.152 0.202 (3.86 5.13)
0.260 0.320 (6.60 8.13) 0.016 0.022 (0.41 0.56) 0.135 0.165 (3.43 4.19) 0.095 0.115 (2.41 2.92) 0.155 0.195 (3.94 4.95)
Y7 0893
15
LT1074/LT1076
U.S. Area Sales Offices
NORTHEAST REGION Linear Technology Corporation One Oxford Valley 2300 E. Lincoln Hwy.,Suite 306 Langhorne, PA 19047 Phone: (215) 757-8578 FAX: (215) 757-5631 Linear Technology Corporation 266 Lowell St., Suite B-8 Wilmington, MA 01887 Phone: (508) 658-3881 FAX: (508) 658-2701 SOUTHEAST REGION Linear Technology Corporation 17060 Dallas Parkway Suite 208 Dallas, TX 75248 Phone: (214) 733-3071 FAX: (214) 380-5138 CENTRAL REGION Linear Technology Corporation Chesapeake Square 229 Mitchell Court, Suite A-25 Addison, IL 60101 Phone: (708) 620-6910 FAX: (708) 620-6977 SOUTHWEST REGION Linear Technology Corporation 22141 Ventura Blvd. Suite 206 Woodland Hills, CA 91364 Phone: (818) 703-0835 FAX: (818) 703-0517 NORTHWEST REGION Linear Technology Corporation 782 Sycamore Dr. Milpitas, CA 95035 Phone: (408) 428-2050 FAX: (408) 432-6331
International Sales Offices
FRANCE Linear Technology S.A.R.L. Immeuble "Le Quartz" 58 Chemin de la Justice 92290 Chatenay Malabry France Phone: 33-1-41079555 FAX: 33-1-46314613 GERMANY Linear Techonolgy GmbH Untere Hauptstr. 9 D-85386 Eching Germany Phone: 49-89-3197410 FAX: 49-89-3194821 JAPAN Linear Technology KK 5F YZ Bldg. 4-4-12 Iidabashi, Chiyoda-Ku Tokyo, 102 Japan Phone: 81-3-3237-7891 FAX: 81-3-3237-8010 KOREA Linear Technology Korea Branch Namsong Building, #505 Itaewon-Dong 260-199 Yongsan-Ku, Seoul Korea Phone: 82-2-792-1617 FAX: 82-2-792-1619 SINGAPORE Linear Technology Pte. Ltd. 101 Boon Keng Road #02-15 Kallang Ind. Estates Singapore 1233 Phone: 65-293-5322 FAX: 65-292-0398 TAIWAN Linear Technology Corporation Rm. 801, No. 46, Sec. 2 Chung Shan N. Rd. Taipei, Taiwan, R.O.C. Phone: 886-2-521-7575 FAX: 886-2-562-2285 UNITED KINGDOM Linear Technology (UK) Ltd. The Coliseum, Riverside Way Camberley, Surrey GU15 3YL United Kingdom Phone: 44-276-677676 FAX: 44-276-64851
World Headquarters
Linear Technology Corporation 1630 McCarthy Blvd. Milpitas, CA 95035-7487 Phone: (408) 432-1900 FAX: (408) 434-0507
0294
16
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7487
(408) 432-1900 q FAX: (408) 434-0507 q TELEX: 499-3977
BA/GP 0494 2K REV B · PRINTED IN USA
© LINEAR TECHNOLOGY CORPORATION 1994