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AN1621 APPLICATION NOTE
300W SECONDARY CONTROLLED TWO-SWITCH FORWARD CONVERTER WITH L5991A
1 INTRODUCTION
A typical off-line isolated switch-mode power supply has the controller located on the primary side of the transformer, whereas the output voltages to be controlled and the housekeeping functions are located on the secondary side. Usually the voltage feedback signal is transferred to the primary controller by using an optocoupler or a transformer. We want to propose here an asymmetrical half bridge forward converter with the controller located on the secondary side. This solution offers some advantages: direct use of the on-board voltage reference no need of the opto feedback, with its temperature and ageing gain dependence available on-board housekeeping functions eliminates the need of an additional dedicated device negligible extra cost on the gate drive transformer to satisfy safety requirements The controller operating on the secondary side requires a specific concept for the start-up sequence, realised here with a very simple, low consumption and cost effective solution.
2 POWER SUPPLY DESCRIPTION
2.1 Topology
Considering the 300W of output power delivered to the load, the most appropriate topology is the asymmetrical half bridge converter. This topology requires two power MOS transistors with a voltage breakdown equal or little higher than the max. rectified mains voltage (thanks to the two clamping diodes), with a proper Rdson to reach the target efficiency. Fig. 1 shows the complete schematic diagram of the 300W Power Supply.
2.2
Start-up circuit
As already mentioned in the introduction section, the controller is located on the secondary side of the power transformer, and for this reason, a start-up circuit has to be provided for a correct system activation. The start-up circuit is based on a diac sending a train of controlled pulses to the low side drive section; the floating drive section is energised by the second secondary gate drive transformer winding . As soon as the L5991a wakes-up, it generates a pwm signal enabling the start-up circuit.
2.3
Gate driver
The two power mosfets, T1 and T2, are driven by a small transformer designed to satisfy the isolation safety requirements and fast switching times. For optimum magnetic coupling (required by the high switching frequency operation) and minimum number of turns, a high permeability core has been selected.
AN1621/1104
Rev. 2 1/14
T1 STW14NK50
D19 1N4148 10T 50T 1T R30 180 D13 STTH1L06 D14 STTH1L06 D15 STTH1L06 D2 IN4148 32T R8 430 D9 18V C16 390pF ISEN R15 10 Tsf2 10T PGND C14 470nF C30 2.2nF Y1 L6 9T D5 1N4148 D3 IN4148 R1 3.6 R2 10 R14 22K C3 100pF D16 1N4148
T5 BC327
10T
D17 1N4148
T4 BC327
T2 STW14NK50
R16 360K
DIAC DB3
VC 13 OUT 9
VCC 8 5
FB 6
COMP
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BYV52-200 10T 1T 77930 C4 1000µF C3 1000µF C2 1000µF C5 1µF P1 10K D18 1N4148 Tsf1 C6 1nF 50T R24 30K D1 L1 24T
220VAC
EMI
AN1621 APPLICATION NOTE
C17 100nF
24V 13A
Vo
C26-27 220µF 400V
R12 6.2K R11 1.2K
R9 3.3K R10 270 C18 100nF
BRIDGE KBUBJ
C15 100nF
R7 80K C29 3 3nF C7 100µF C9 1nF
Figure 1. Schematic Diagram of 300W Power Supply
R23 180
R19 30K
DIS 14
R20 7.5K
10 11 SGND 12 SS 7
L5991A
2 RCT C10 2.2nF VREF R5 4.7K 4 DCL 15
16
ST-BY
T3 BC337
C25 100nF IN455_MOD
C28 330pF
D12 1N4148
C12 1nF
C11 4.7µF
AN1621 APPLICATION NOTE Table 1. Power Supply Specification
Symbol VI VO IO Il Po fs Description Input voltage Output voltage Output Current Limiting Current Output continuos power Switching Frequency Target Efficiency (full load) Parameter 220Vac (176 Vac to 265 Vac) // 50Hz 24V % (ripple voltage <1%) 13Amax. continuous, 0.5Amin constant type till ground 312W 200kHz = 90% (from mains to output)
The core is E20/10/5-3C85, 10 turns/winding, no air-gap. A high permeability core minimises the magnetising current to maintain a correct operation far away of the core saturation point, at maximum duty cycle and high core temperature. The proposed gate drive circuit allows the use of the 1:1:1 turns ratio drive transformer; moreover, a controlled drain current rise time (by R15) and a fast fall time are achieved. A very short min. Ton pulse give the possibility to stabilise the output voltage at max. mains and min. load.
2.4
Power MOS selection
The two-switch topology allows to use the power elements with a voltage breakdown equal to the max. rectified mains voltage. The mosfet used here is the STW14NK50; this device, with 500V of BVdss give us also some safety margin. The basic parameters of the STW14NK50 are listed below: Rdson (25°C) = 0.38 max., at Id = 7A (0.76 max. at 100°C) Coss = 300pF typ., Qg = 75nC typ., package in TO-247 At min. supply voltage and max. load current, the conduction losses for each transistor are: Pcon = I2rmsp · Rdson(100°C) = 2.232 · 0.76 = 3.8W where the effective Irmsp2 is calculated in the power transformer section. Estimating in about 3.2W switching and parasitic losses, the total power losses of each transistor are about 7W. Considering 100°C of maximum operating junction temperature (at 40°C of ambient temperature) and a thermal resistance junction-heatsink of 0.66°C/W, a heatsink of 4°C/W is required to dissipate both the transistors.
2.5
Current sense
Considering the output current rating of 13A continuous, it's our opinion that a current transformer for current sensing is the best approach for maximising the efficiency, reliability and internal ambient temperature in case the power supply has to be housed in a plastic box. Due to the constant current limiting requirements, as shown in Fig2, a couple of current transformers have been used; one transformer is sensing the current flowing into D1 (in conduction when T1 and T2 are ON) and the second one is sensing the current flowing into D5, recirculation diode. Oring the two transformers by D2 and D3, and closing the loop with a proper impedance value, R1, we realise a voltage signal reproducing exactly the inductor current shape. The current sensing loop is closed by R1 selected according the transformers turns. Two small toroid ferrite cores (41005-TC, Magnetics, F material, 3000µ) have been used, with 50 turns. R1 is defined by: 50 1V R1 = -----------------lpk where: 1V is the nominal threshold voltage of the current sense. Ipk is the inductor peak current( considering a 20% of current ripple, Ipk = Io + I/2 = 13 + 1.3 = 14.3A)
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AN1621 APPLICATION NOTE Figure 2. Output Current Limiting Characteristic Using Two Current Transformer
Vo 25 20 15 10 5 0
D96IN456
8
10
12
14
Io
The calculated value is R1=3.5 Fig. 2 shows the constant current characteristic using two current transformers. The difference from the two current values, at output short-circuit and at current limiting intervention, is proportional to the half of chocke ripple current.A choke with higher value or higher switching frequency, can reduce this difference. If constant current feature is not requested to be constant till the output is reaching zero V, one single current transformer can be used. The new limiting current characteristic and the schematic diagram are shown in Fig 3a and 3b:In order to reduce the peak current before hiccup intervention, an offset can be superimposed to the current sensing circuit to anticipate the hiccup limiting current intervention, by using the additional network shown below ( figs. 4a and 4b): Using this solution, also the I/Vo is higher, reducing the difference from the intervention point and the short circuit current values.This solution can be used with two current transformers too.
Figure 3. Output Current Limiting Characteristic Using One Current Transformer
Vo 25 20 15 10 5 0
D2 R8 430 R1 3.5
D96IN457
1T C6 10T R2
D1
L1
50T
D5
C16 330pF 13
ISEN
4
6
8
10
12
14 Io
L5991A
D96IN458_mod
a: Output current limiting characteristic using one current transformer.
b: limiting current schematic diagram
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AN1621 APPLICATION NOTE Figure 4. Output Current Limiting Characteristic with Foldback
1T D1 L1 P1
Vo 25
D96IN459
50T C6 10T R2 D5 C4 C3 C2
3K
3K
20
R11 1.2K
15
D2 R1 120K
10
R8
5
ISEN
C16
2K VREF
0
13
4
4
6
8
10
12
14 Io
D96IN460_mod
L5991A
5
FB
a: Output current limiting characteristic
b: Schematic diagram of the modified hiccup limiting current thereshold.
2.6
Output Diode selection
The reverse voltage of the output diodes is given by the formula: Vr = Vin max/n = 375/3.3= 114V, where n is the transformer turns ratio. For a correct functionality a ultra-fast recovery diode is requested, mainly to limit switching losses and EMI problems. To calculate the conduction losses the following equation has been applied to the selected type, BYV52-200. For the single diode the formula is: P = 0.7 · I(AV) + 0.0075 · I2(RMS) that, rearranged to take into account both the diodes, becomes: P = 0.7 · IOMAX + 0.0075 · IO2MAX = 10.4W where IOMAX = 13A. Considering the TO247 package (1.2 °C/W total thermal resistance junction-heatsink), to ensure that the junction temperature does not exceed 100 °C at 40 °C max. ambient temperature, the heatsink has to be dimensioned for about 4°C/W.
2.7
Power transformer design
The forward transformer delivers energy from the primary to the secondary without any storage. The only consideration is with regards of the magnetising current, that has to be limited at a safety value far from core saturation. Our core selection is based on AP, area product, defined as AP = Aw · Ae.
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AN1621 APPLICATION NOTE
The minimum AP to avoid saturation is: 67.2 P O 1.31 4 AP = ----------------------- cm B f The AP necessary to limit the core temperature rise not above 30 °C is 235 P O 1.58 2 0.66 4 AP = --------------------- ( Kh f + Ke f ) cm f Where: Po = output power B = flux swing f = switching frequency = efficiency Ke =4 · 10-10 Kh = 4 · 10-5 . In this application (Po = 312W, = 90%, f = 200KHz) and considering a maximum B = 200mT for saturation, the most stringent condition is determined by the formula related to the core temperature raise. The minimum AP required is 1.97 cm4 . The selected core is ETD39 (AP = 2.2 cm4 ) in 3F3 material. Assuming total losses of 1% of Po (3W), 2/3 (2W) for the core and 1/3 (1W) for the copper, the specific core losses are:
3 Pfe 2W P = --------- = ---------------------- = 0.173W/cm 3 Ve 11.5cm
In presence of such a power losses, the flux swing can be determined, for 3F3 material, by using the diagram of material, at 200KHz :
Figure 5. Specific Power Loss as a Function of Frequency Peak Flux Density with Frequency as a Parameter
Ptot (KW/m3) 5K 2K 1K
100
D96IN461
Tamb=100°C
700 KH z
3F3
500
Hz
200 100 50 20 10 1 2 5
1M
KH
z
200 KH z
25K Hz
400
KHz
10 20
50 100 200
B(mT)
or by using the empirical formula
0.416 0.416 P 0.173 B = ------------------------------------ = ----------------------------------------------------------------------------------------------------------------- = 130mT 2 4 10 5 200 10 3 + 4 10 10 ( 200 10 3 ) 2 Kh f + Ke f
With a specific core losses of 173 mW/cm3 the flux swing must not exceed 130mT The minimum primary turns is given by:
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AN1621 APPLICATION NOTE
V inmindc T on ( max ) 4 200 2.4 N P -------------------------------------------------- 10 = ---------------------------- = 29.5 0.130 125 B A e where Ton is in sec, B in Tesla and Ae in mm2. The turns ratio is defined as: V in ( min ) D max 200 0.48 n = 0.9 --------------------------------------- = 0.9 ------------------------------ = 3.38 Vo + Vf + Vp 24 + 1 + 0.5 where: Dmax = 0.48 (maximum duty cycle): Vf =1V (voltage drop of the output diode); Vp = 0.5V (voltage drop of the output inductor). The primary and secondary turns number are Np = 32, Ns = 10. As for the wire selection, the skin effect is not negligible at this frequency. To reduce this effect, the wire diameter should not exceed two times the penetration depth: 7.5 = ------- = 0.17mm f and therefore a wire diameter of 0.36mm (AWG27) is suitable. However, to have a sufficient copper area, a certain number of these wires will be twisted together. To define the numbers of wires this equation can be used: I rms N wire = R N turns I w ------------P loss where: R = resistance of the wire per cm lw = winding length (cm) As to the primary side: Po 312 I rmsp = ----------------------------------------------------- = ------------------------------------------ = 2.23A V inmindc D max 0.9 200 0.48 The RMS secondary current is 3.4 times the primary one. Considering the copper losses equally dissipated between primary and secondary windings, this yields 3 wires in parallel (Npwire) for the primary winding and 10 wires (Nswire) for secondary winding. The ETD39 core has a winding area of 1.7 cm2 but the real available space is approximately 40% (0.68 cm2): isolation requirements and fill factor must be taken into account. The total copper area is: Atot = Ais · (Npwire · Np + Nswire · Ns) = 0.255 cm2 that fits the window. The windings are interleaved: one layer of 16 turns (first half of the primary), one layer of 10 turns (the secondary) and another layer of 16 turns (second half of the primary). With this arrangement the transformer used in the application has a leakage inductance of 15uH (about 0.5% of the primary inductance, which is 2.7 mH). The magnetising current is:
6 V in ( min ) t on 200 2.4 10 I m = -------------------------------- = -------------------------------------- = 180mA 3 Lp 2.7 10 2
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AN1621 APPLICATION NOTE
The primary peak current is: Po 312 I pk = ------------------------------------------------- = -------------------------------------- = 3.22A V inmindc D max 0.9 200 0.48 A good condition is when Im 10% of Ipk; in this case Im is imposed lower than 6% of Ip.
2.8
Output inductor & auxiliary supply
To calculate the inductor value, the max. current ripple has been set at 2.6A, 20% of Iomax. The max. current ripple occurs at max. input voltage; in this operating condition the min duty cycle is: n ( V o + V loss ) 3.2 ( 24 + 1.5 ) D min = --------------------------------------- = -------------------------------------- = 0.22 375 V inmax The inductor value is defined taking into account the current downslope which corresponds to the off-time of the power switches: Toff max. = T · (1-Dmin) = 5 · (1-0.22) = 3.9µs that yields:
6 ( V o + V loss ) T offmax ( 24 + 1.5 ) 3.9 10 L o = -------------------------------------------------------- = ------------------------------------------------------- = 39µH 2.6 I max
To realise the inductor, a Magnetic's Kool Mm Core 77930 has been used. Since the inductor has to assure the requested value at maximum load current, the turns number must be calculated taking into account the roll-off of the initial permeability. This results in 24 turns, that leads to 90µH at zero load. We can take advantage of this non-linear characteristic to keep a good output regulation and stability when working at min. load current( around 0.5A for this case).From the output inductor it is drown also the energy necessary to supply the L5991A controller by introducing a 9 turns auxiliary winding.
2.9
Input capacitor
The input filtering capacitor has to be dimensioned in order to deliver the requested max. output power, at min. mains value, with a reasonable ripple voltage at 100Hz. The design process has to consider also the rms current flowing into the capacitor ( a major reason of stress) and the voltage rating. The minimum capacitor value is defined by: Po 312 C = ------------------------------------------------------- = ----------------------------------------------------------- = 310µF 2 2 2 2 F 50 ( V Cp V Cm ) 0.9 50 ( 248 200 ) where: Vcp is the peak value at minimum mains voltage, 248V Vcm is the minimum value at minimum mains voltage, 200V The chosen capacitors are 2 x 220µF - 400V EYS 06. At this point the new Vcm value is 215V. The conduction time is:
1 215 ---------- cos --------- 248 V cp 3 t c = ------------------------------- = ------------------------------ = 1.68 10 sec 2 3.14 50 2 F 50
cos
1 V cm
8/14
AN1621 APPLICATION NOTE
The input capacitor peak current is:
6 C ( V cp V cm ) 440 10 ( 248 215 ) I ch = --------------------------------------- = ------------------------------------------------------------- = 8.8A 3 tc 1.6 10
The RMS current that the capacitor has to substain is: I rms = I ch 2 t c F 50 ( 2 t c F 50 ) = 8.8 3.2 10 I rms200kHz 2 I rms50 + --------------------------- = 1.6
2 2 3
50 ( 3.2 10
3
50 )
2
= 3.32A
I rmstot =
3.32 + 2.23
2
2
= 3.76A
Considering the sensitivity to the temperature of the electrolitic capacitors, and the difficulties in fixing exactly the effective rms current value, for this particular part number, with an ambient temperature of 40°C, the permitted temperature case is 85 °C. The selected capacitors show a 1.4Arms each of current capability at Tcase = 85°C; that's enough to sustain our operating conditions.
2.10 Output capacitors
The output capacitor value is selected on the basis of the output voltage ripple requirement. This ripple is basically due to the ESR, since the capacitive component is by far lower. Then it must be ensured that the total ESR is below a maximum value of: V ripple 0.24V ESR max = -------------------- = --------------- = 0.092 I max 2.6A where the ripple voltage has been fixed at 1% max. of Vo. Three capacitors EKE 1000µF/35V (ROE) have been paralleled, for a total ESR of about 23mOhm.
2.11 Compensation network
The power supply system can be simplified in two parts: Power and Feedback loop blocks.
Figure 6. Closed Loop Block Diagram
INT VREF (2.5V)
+
+ -
V'O
GE/A
VC
GPW
VO
P
D96IN462
The transfer function of the Power block is: S 1 + ----Sz V o n G pw ( s ) = --------- = -------------- R o ---------------V c 3 Rs S 1 + -----Sp is the small signal output voltage is the small signal error amplifier output voltage
where: Vo Vc
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AN1621 APPLICATION NOTE
Sp is determined by the load (Ro) and the output capacitance Sz is determined by the ESR and the value of the output capacitance n is the current transformer turn number Rs is the sense resistance A load regulation of 1.5% has been imposed that corresponding to a maximum output voltage variation of 360mV. The power block DC gain is: V o n Ro G pw = --------- = -------------3 Rs V c Considering a load variation between Io=0.1A and Io=13A, that corresponds to Ro = 240ohm and Ro = 1.84, the Vc excursion is: Vo 3 Rs 1 1 V c = ------------------------- --------------- ---------------- = 2.7V R R omax n omin The E/A DC gain and the voltage divisor must be selected to fit this condition V o --------- = 7.5 V c The output voltage divider usually sinks a current of about 2mA and the resistance values are: R11 = 12k P1+ R12 = 1k The R7 value is: V o R7 = --------- ( P1 + R12 ) = 80k V c It is necessary to introduce a pole at about one third of the zero frequency and C9 is: 1 C9 = --------------------- = 1.8nF 2 Fz --------------------3 R7
2.12 Overvoltage protection
The voltage divider, R9 and R10, is providing for fixing the threshold for overvoltage protection intervention. With the selected values, the threshold is fixed at 30V. C18 is fixing the min. overvoltage time intervention to avoid OVP triggering from spikes. For an OVP function in tracking with the output voltage, in particular when Vo is adjustable, it's enough to substitute the R11 resistor with a voltage divider, and to connect the common point to pin14.
2.13 Evaluation results
Fig 7 show the efficiency values obtained in different working conditions of input voltages and output currents. The efficiency is from ac mains to output dc voltage, at a measured switching frequency of 200kHz.
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AN1621 APPLICATION NOTE Figure 7. Efficiency versus Output Load Current at min.,typ. and max. AC Voltage.
(%) 90 85 80 75 70 65 0 2 4 6 8 10 12 14 265Vac
D96IN463
176Vac 220Vac
The efficiency is quite constant and equal or above 85% from 2A to full load. At min. load of 0.5A, it's 70% at 220Vac. The total losses, for an amount of 3.6W are mainly due to Coss of the power mos devices and transformer core losses due to the magnetizing current. Fig 8. shows the load transient response, at nominal ac input voltage value and output load variation from 1A to 12A, in a couple of msec, and fig 9 shows the enlarged section at the moment of current load rise. Fig 10. To be noticed the absence of output overshoot on the output voltage.
Figure 8. Output Voltage Response to Load Transient
D96IN464
CH1: Vo (100mV/div)
CH2: Io (5A/div)
T: 1ms/div
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AN1621 APPLICATION NOTE Figure 9. Enlarged Section of Load Transient Response at Load Current Rise
D96IN465
CH2: Io (2A/div)
CH1: Vo (100mV/div)
T:100µs/div
Figure 10. Turn-on at Mains Switch-on
IN466_mod
CH1: Vo (10V/div)
CH2: L5991A Self Supply Voltage C7=47µF (5V/div)
T:100ms/div
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AN1621 APPLICATION NOTE Table 2. Revision History
Date November 2002 November 2004 Revision 1 2 First Issue in EDOCD New Style-sheet in compliance to the "Corporate Technical Pubblications Desin Guide" and labeling of Schematic Diagram Description of Changes
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AN1621 APPLICATION NOTE
The present note which is for guidance only, aims at providing customers with information regarding their products in order for them to save time. As a result, STMicroelectronics shall not be held liable for any direct, indirect or consequential damages with respect to any claims arising from the content of such a note and/or the use made by customers of the information contained herein in connection with their products.
Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications mentioned in this publication are subject to change without notice. This publication supersedes and replaces all information previously supplied. STMicroelectronics products are not authorized for use as critical components in life support devices or systems without express written approval of STMicroelectronics. The ST logo is a registered trademark of STMicroelectronics. All other names are the property of their respective owners © 2004 STMicroelectronics - All rights reserved STMicroelectronics group of companies Australia - Belgium - Brazil - Canada - China - Czech Republic - Finland - France - Germany - Hong Kong - India - Israel - Italy - Japan Malaysia - Malta - Morocco - Singapore - Spain - Sweden - Switzerland - United Kingdom - United States of America www.st.com
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